Differential frequency down-conversion using techniques of universal frequency translation technology

ABSTRACT

Methods, systems, and apparatuses for down-converting an electromagnetic (EM) signal by aliasing the EM signal, and applications thereof are described herein. Reducing or eliminating DC offset voltages and re-radiation generated when down-converting an electromagnetic (EM) signal is also described herein. Down-converting a signal and improving receiver dynamic range is also described herein.

[0001] This application claims priority to the following: U.S.Provisional Application No. 60/171,502, filed Dec. 22, 1999; U.S.Provisional Application No. 60/177,705, filed on Jan. 24, 2000; U.S.Provisional Application No. 60/129,839, filed on Apr. 16, 1999; U.S.Provisional Application No. 60/158,047, filed on Oct. 7, 1999; U.S.Provisional Application No. 60/171,349, filed on Dec. 21, 1999; U.S.Provisional Application No. 60/177,702, filed on Jan. 24, 2000; U.S.Provisional Application No. 60/180,667, filed on Feb. 7, 2000; and U.S.Provisional Application No. 60/171,496, filed on Dec. 22, 1999, all ofwhich are incorporated by reference herein in their entireties.

CROSS-REFERENCE TO OTHER APPLICATIONS

[0002] The following applications of common assignee are related to thepresent application, and are herein incorporated by reference in theirentireties:

[0003] “Method and System for Down-Converting Electromagnetic Signals,”Ser. No. 09/176,022, filed Oct. 21, 1998.

[0004] “Method and System for Frequency Up-Conversion,” Ser. No.09/176,154, filed Oct. 21, 1998.

[0005] “Method and System for Ensuring Reception of a CommunicationsSignal,” Ser. No. 09/176,415, filed Oct. 21, 1998.

[0006] “Integrated Frequency Translation And Selectivity,” Ser. No.09/175,966, filed Oct. 21, 1998.

[0007] “Applications of Universal Frequency Translation,” Ser. No.09/261,129, filed Mar. 3, 1999.

[0008] “Method and System for Down-Converting Electromagnetic SignalsHaving Optimized Switch Structures,” Ser. No. 09/293,095, filed Apr. 16,1999.

[0009] “Method and System for Down-Converting Electromagnetic SignalsIncluding Resonant Structures for Enhanced Energy Transfer,” Ser. No.09/293,342, filed Apr. 16, 1999.

[0010] “Method and System for Frequency Up-Conversion with a Variety ofTransmitter Configurations,” Ser. No. 09/293,580, filed Apr. 16, 1999.

[0011] “Integrated Frequency Translation and Selectivity with a Varietyof Filter Embodiments,” Ser. No. 09/293,283, filed Apr. 16, 1999.

[0012] “Matched Filter Characterization and Implementation of UniversalFrequency Translation Method and Apparatus,” Ser. No. ______, Atty.Docket No. 1744.0920000, filed Mar. 9, 2000.

BACKGROUND OF THE INVENTION

[0013] 1. Field of the Invention

[0014] The present invention relates to down-conversion andup-conversion of electromagnetic (EM) signals. More particularly, thepresent invention relates to reducing or eliminating DC offset voltagesand re-radiation that occurs during down-conversion of EM signals tointermediate frequency or baseband signals.

[0015] 2. Related Art

[0016] Electromagnetic (EM) information signals (baseband signals)include, but are not limited to, video baseband signals, voice basebandsignals, computer baseband signals, etc. Baseband signals include analogbaseband signals and digital baseband signals.

[0017] It is often beneficial to propagate EM signals at higherfrequencies. This is generally true regardless of whether thepropagation medium is wire, optic fiber, space, air, liquid, etc. Toenhance efficiency and practicality, such as improved ability to radiateand added ability for multiple channels of baseband signals,up-conversion to a higher frequency is utilized. Conventionalup-conversion processes modulate higher frequency carrier signals withbaseband signals. Modulation refers to a variety of techniques forimpressing information from the baseband signals onto the higherfrequency carrier signals. The resultant signals are referred to hereinas modulated carrier signals. For example, the amplitude of an AMcarrier signal varies in relation to changes in the baseband signal, thefrequency of an FM carrier signal varies in relation to changes in thebaseband signal, and the phase of a PM carrier signal varies in relationto changes in the baseband signal.

[0018] In order to process the information that was in the basebandsignal, the information must be extracted, or demodulated, from themodulated carrier signal. However, because conventional signalprocessing technology is limited in operational speed, conventionalsignal processing technology cannot easily demodulate a baseband signalfrom higher frequency modulated carrier signal directly. Instead, higherfrequency modulated carrier signals must be down-converted to anintermediate frequency (IF), from where a conventional demodulator candemodulate the baseband signal.

[0019] Conventional down-converters include electrical components whoseproperties are frequency dependent. As a result, conventionaldown-converters are designed around specific frequencies or frequencyranges and do not work well outside their designed frequency range.

[0020] Conventional down-converters generate unwanted image signals andthus must include filters for filtering the unwanted image signals.However, such filters reduce the power level of the modulated carriersignals. As a result, conventional down-converters include poweramplifiers, which require external energy sources.

[0021] When a received modulated carrier signal is relatively weak, asin, for example, a radio receiver, conventional down-converters includeadditional power amplifiers, which require additional external energy.

SUMMARY OF THE INVENTION

[0022] Briefly stated, the present invention is directed to methods,systems, and apparatuses for down-converting an electromagnetic (EM)signal by aliasing the EM signal, and applications thereof. The presentinvention is further directed to reducing or eliminating DC offsetvoltages and re-radiation generated when down-converting anelectromagnetic (EM) signal. The present invention is still furtherdirected to improving receiver dynamic range.

[0023] Generally, the invention operates by receiving an EM signal. Theinvention also receives an aliasing signal having an aliasing rate. Theinvention aliases the EM signal according to the aliasing signal todown-convert the EM signal.

[0024] In an embodiment, the invention down-converts the EM signal to anintermediate frequency (IF) signal.

[0025] In another embodiment, the invention down-converts the EM signalto a demodulated baseband information signal.

[0026] In another embodiment, the FM signal is a frequency modulated(FM) signal, which is down-converted to a non-FM signal, such as a phasemodulated (PM) signal or an amplitude modulated (AM) signal.

[0027] In another embodiment, the EM signal is an I/Q modulated signal,which is down-converted to an in-phase information signal and aquadrature-phase information signal.

[0028] The invention is applicable to any type of EM signal, includingbut not limited to, modulated carrier signals (the invention isapplicable to any modulation scheme or combination thereof) andunmodulated carrier signals.

[0029] Further features and advantages of the invention, as well as thestructure and operation of various embodiments of the invention, aredescribed in detail below with reference to the accompanying drawings.In the drawings, like reference numbers generally indicate identical,functionally similar, and/or structurally similar elements. The drawingin which an element first appears is generally indicated by theleft-most digit(s) in the corresponding reference number.

BRIEF DESCRIPTION OF THE FIGURES

[0030] The invention shall be described with reference to theaccompanying figures, wherein:

[0031]FIG. 1A is a block diagram of a universal frequency translation(UFT) module according to an embodiment of the invention.

[0032]FIG. 1B is a more detailed diagram of a universal frequencytranslation (UFT) module according to an embodiment of the invention.

[0033]FIG. 1C illustrates a UFT module used in a universal frequencydown-conversion (UFD) module according to an embodiment of theinvention.

[0034]FIG. 1D illustrates a UFT module used in a universal frequencyup-conversion (UFU) module according to an embodiment of the invention.

[0035]FIG. 2 is a block diagram of a universal frequency translation(UFT) module according to an alternative embodiment of the invention.

[0036]FIG. 3 is a block diagram of a universal frequency up-conversion(UFU) module according to an embodiment of the invention.

[0037]FIG. 4 is a more detailed diagram of a universal frequencyup-conversion (UFU) module according to an embodiment of the invention.

[0038]FIG. 5 is a block diagram of a universal frequency up-conversion(UFU) module according to an alternative embodiment of the invention.

[0039] FIGS. 6A-6I illustrate example waveforms used to describe theoperation of the UFU module.

[0040]FIG. 7 illustrates a UFT module used in a receiver according to anembodiment of the invention.

[0041]FIG. 8 illustrates a UFT module used in a transmitter according toan embodiment of the invention.

[0042]FIG. 9 illustrates an environment comprising a transmitter and areceiver, each of which may be implemented using a UFT module of theinvention.

[0043]FIG. 10 illustrates a transceiver according to an embodiment ofthe invention.

[0044]FIG. 11 illustrates a transceiver according to an alternativeembodiment of the invention.

[0045]FIG. 12 illustrates an environment comprising a transmitter and areceiver, each of which may be implemented using enhanced signalreception (ESR) components of the invention.

[0046]FIG. 13 illustrates a UFT module used in a unified down-conversionand filtering (UDF) module according to an embodiment of the invention.

[0047]FIG. 14 illustrates an example receiver implemented using a UDFmodule according to an embodiment of the invention.

[0048] FIGS. 15A-15F illustrate example applications of the UDF moduleaccording to embodiments of the invention.

[0049]FIG. 16 illustrates an environment comprising a transmitter and areceiver, each of which may be implemented using enhanced signalreception (ESR) components of the invention, wherein the receiver may befurther implemented using one or more UFD modules of the invention.

[0050]FIG. 17 illustrates a unified down-converting and filtering (UDF)module according to an embodiment of the invention.

[0051]FIG. 18 is a table of example values at nodes in the UDF module ofFIG. 17.

[0052]FIG. 19 is a detailed diagram of an example UDF module accordingto an embodiment of the invention.

[0053]FIGS. 20A and 20G are example aliasing modules according toembodiments of the invention.

[0054] FIGS. 20B-20F are example waveforms used to describe theoperation of the aliasing modules of FIGS. 20A and 20G.

[0055]FIG. 21 illustrates an enhanced signal reception system accordingto an embodiment of the invention.

[0056] FIGS. 22A-22F are example waveforms used to describe the systemof FIG. 21.

[0057]FIG. 23A illustrates an example transmitter in an enhanced signalreception system according to an embodiment of the invention.

[0058]FIGS. 23B and 23C are example waveforms used to further describethe enhanced signal reception system according to an embodiment of theinvention.

[0059]FIG. 23D illustrates another example transmitter in an enhancedsignal reception system according to an embodiment of the invention.

[0060]FIGS. 23E and 23F are example waveforms used to further describethe enhanced signal reception system according to an embodiment of theinvention.

[0061]FIG. 24A illustrates an example receiver in an enhanced signalreception system according to an embodiment of the invention.

[0062] FIGS. 24B-24J are example waveforms used to further describe theenhanced signal reception system according to an embodiment of theinvention.

[0063]FIG. 25 illustrates an exemplary I/Q modulation receiver,according to an embodiment of the present invention.

[0064]FIG. 26 illustrates a I/Q modulation control signal generator,according to an embodiment of the present invention.

[0065]FIG. 27 illustrates example waveforms related to the I/Qmodulation control signal generator of FIG. 26.

[0066]FIG. 28 illustrates example control signal waveforms overlaid uponan input RF signal.

[0067]FIG. 29 illustrates a I/Q modulation receiver circuit diagram,according to an embodiment of the present invention.

[0068] FIGS. 30-40 illustrate example waveforms related to the receiverof FIG. 29.

[0069]FIG. 41 illustrates a single channel receiver, according to anembodiment of the present invention.

[0070]FIG. 42 illustrates an alternative I/Q modulation receiver,according to an embodiment of the present invention.

[0071]FIG. 43 illustrates an I/Q modulation transmitter, according to anembodiment of the present invention.

[0072]FIG. 44A illustrates an example antenna that transmitsre-radiation.

[0073] FIGS. 44B-D illustrates example signals and frequency spectrumsrelated to re-radiation effects.

[0074] FIGS. 45A-D illustrate example implementations of a switch moduleaccording to embodiments of the invention.

[0075] FIGS. 46A-D illustrate example aperture generators.

[0076]FIG. 46E illustrates an oscillator according to an embodiment ofthe present invention.

[0077]FIG. 47 illustrates an energy transfer system with an optionalenergy transfer signal module according to an embodiment of theinvention.

[0078]FIG. 48 illustrates an aliasing module with input and outputimpedance match according to an embodiment of the invention.

[0079]FIG. 49A illustrates an example pulse generator.

[0080]FIGS. 49B and C illustrate example waveforms related to the pulsegenerator of FIG. 49A.

[0081]FIG. 50 illustrates an example energy transfer module with aswitch module and a reactive storage module according to an embodimentof the invention.

[0082] FIGS. 51A-B illustrate example energy transfer systems accordingto embodiments of the invention.

[0083]FIG. 52A illustrates an example energy transfer signal moduleaccording to an embodiment of the present invention.

[0084]FIG. 52B illustrates a flowchart of state machine operationaccording to an embodiment of the present invention.

[0085]FIG. 52C is an example energy transfer signal module.

[0086]FIG. 53 is a schematic diagram of a circuit to down-convert a 915MHZ signal to a 5 MHZ signal using a 101.1 MHZ clock according to anembodiment of the present invention.

[0087]FIG. 54 shows example simulation waveforms for the circuit of FIG.53 according to embodiments of the present invention.

[0088]FIG. 55 is a schematic diagram of a circuit to down-convert a 915MHZ signal to a 5 MHZ signal using a 101 MHZ clock according to anembodiment of the present invention.

[0089]FIG. 56 shows example simulation waveforms for the circuit of FIG.55 according to embodiments of the present invention.

[0090]FIG. 57 is a schematic diagram of a circuit to down-convert a 915MHZ signal to a 5 MHZ signal using a 101.1 MHZ clock according to anembodiment of the present invention.

[0091]FIG. 58 shows example simulation waveforms for the circuit of FIG.57 according to an embodiment of the present invention.

[0092]FIG. 59 shows a schematic of the circuit in FIG. 53 connected toan FSK source that alternates between 913 and 917 MHZ at a baud rate of500 Kbaud according to an embodiment of the present invention.

[0093]FIG. 60A illustrates an example energy transfer system accordingto an embodiment of the invention.

[0094] FIGS. 60B-C illustrate example timing diagrams for the examplesystem of FIG. 60A.

[0095]FIG. 61 illustrates an example bypass network according to anembodiment of the invention.

[0096]FIG. 62 illustrates an example bypass network according to anembodiment of the invention.

[0097]FIG. 63 illustrates an example embodiment of the invention.

[0098]FIG. 64A illustrates an example real time aperture control circuitaccording to an embodiment of the invention.

[0099]FIG. 64B illustrates a timing diagram of an example clock signalfor real time aperture control, according to an embodiment of theinvention.

[0100]FIG. 64C illustrates a timing diagram of an example optionalenable signal for real time aperture control, according to an embodimentof the invention.

[0101]FIG. 64D illustrates a timing diagram of an inverted clock signalfor real time aperture control, according to an embodiment of theinvention.

[0102]FIG. 64E illustrates a timing diagram of an example delayed clocksignal for real time aperture control, according to an embodiment of theinvention.

[0103]FIG. 64F illustrates a timing diagram ofan example energy transferincluding pulses having apertures that are controlled in real time,according to an embodiment of the invention.

[0104]FIG. 65 illustrates an example embodiment of the invention.

[0105]FIG. 66 illustrates an example embodiment of the invention.

[0106]FIG. 67 illustrates an example embodiment of the invention.

[0107]FIG. 68 illustrates an example embodiment of the invention.

[0108]FIG. 69A is a timing diagram for the example embodiment of FIG.65.

[0109]FIG. 69B is a timing diagram for the example embodiment of FIG.66.

[0110]FIG. 70A is a timing diagram for the example embodiment of FIG.67.

[0111]FIG. 70B is a timing diagram for the example embodiment of FIG.68.

[0112]FIG. 71A illustrates and example embodiment of the invention.

[0113]FIG. 71B illustrates example equations for determining chargetransfer, in accordance with the present invention.

[0114]FIG. 71C illustrates relationships between capacitor charging andaperture, in accordance with an embodiment of the present invention.

[0115]FIG. 71D illustrates relationships between capacitor charging andaperture, in accordance with an embodiment of the present invention.

[0116]FIG. 71E illustrates power-charge relationship equations, inaccordance with an embodiment of the present invention.

[0117]FIG. 71F illustrates insertion loss equations, in accordance withan embodiment of the present invention.

[0118]FIG. 72 shows the original FSK waveform 5902 and thedown-converted waveform 5904.

[0119]FIG. 73 illustrates a down-converter according to an embodiment ofthe present invention, showing some DC offset contributions.

[0120]FIG. 74 illustrates a down-converter according to an embodiment ofthe present invention, that removes at least some DC offsetcontributions.

[0121]FIGS. 75 and 76 illustrate circuit diagrams according toembodiments of the present invention.

[0122]FIG. 77A illustrates an example clock pulse train.

[0123]FIG. 77B illustrates an example clock frequency spectrum.

[0124]FIG. 78 illustrates a circuit diagram according to an embodimentof the present invention, which may be used to measure DC offsets.

[0125]FIGS. 79 and 80 illustrate example output offset plots for thecircuit diagram of FIG. 78, for a variety of clock signals.

[0126]FIGS. 81 and 82 show example output offset plots obtained for thecircuit model of FIG. 78, with variations in the bond wire inductance.

[0127]FIG. 83 illustrates example V_(OCI) response for a variety ofclock signal rise and fall times.

[0128]FIGS. 84A, 84B, 85A and 85B show the results on an I port of anI/Q receiver according to an embodiment of the present invention, for avariety of LO drive levels and 3 operating channels, for two differentassemblies.

[0129]FIG. 86A illustrates example complimentary architecture outputoffset for a variety of clock signal pulse widths.

[0130]FIG. 86B shows an example spectral plot of a carrier tone at RF,corresponding to LO re-radiation at a UFD module.

[0131]FIG. 86C illustrates the LO re-radiation spectrum shown in FIG.86B after modulation by an example modified maximal length linear PNsequence.

[0132]FIG. 86D shows an example PN modulated output of a UFD module.

[0133]FIG. 86E illustrates the result of FIG. 86D after PN rectificationor correlation.

[0134]FIG. 86F illustrates the result of FIG. 86E after low passfiltering to recover the baseband beat note.

[0135]FIG. 86G illustrates an exemplary signal input harmonic spectrumand conversion clock harmonic spectrum.

[0136]FIG. 86H illustrates an exemplary power series.

[0137]FIG. 861 illustrates an exemplary system block diagram, accordingto an embodiment of the present invention.

[0138]FIG. 87 shows a conventional wireless communicationsdown-conversion system.

[0139]FIG. 88A shows an exemplary down-conversion system that reducesoutput DC offset, according to an embodiment of the present invention.

[0140] FIGS. 88B-H show example waveforms related to the system of FIG.88A, according to an embodiment of the present invention.

[0141]FIG. 89 shows an exemplary down-conversion system that reducesoutput DC offset, according to an embodiment of the present invention.

[0142]FIG. 90 illustrates some aspects of charge injection related tothe present invention.

[0143]FIG. 91 illustrates an exemplary circuit configuration forreducing DC offset voltage caused by charge injection, according to anembodiment of the present invention.

[0144]FIG. 92A illustrates an exemplary down-conversion system,according to an embodiment of the present invention, that may be used toindicate potential points in a signal path where DC offset voltages maybe injected.

[0145]FIG. 92B illustrates an exemplary auto-zero compensation circuitfor reducing or eliminating DC offset inserted by circuit components,according to an embodiment of the present invention.

[0146]FIG. 93 illustrates an exemplary differential DC offset voltagecancellation circuit, according to an embodiment of the presentinvention.

[0147]FIG. 94A illustrates a second exemplary differential DC offsetvoltage cancellation circuit, according to an embodiment of the presentinvention.

[0148] FIGS. 94B-H illustrate example waveforms related to the circuitof FIG. 94A, according to an embodiment of the present invention.

[0149]FIG. 95 illustrates an exemplary differential receiver circuit,according to an embodiment of the present invention.

[0150]FIG. 96 illustrates an exemplary input RF signal and exemplarycontrol signal waveforms, according to embodiments of the presentinvention.

[0151]FIG. 97 illustrates an exemplary I/Q modulation receiver circuit,according to an embodiment of the present invention.

[0152] FIGS. 98A-981 show an exemplary input RF I/Q signal, and severalexemplary control signal waveforms.

[0153]FIG. 99 illustrates an exemplary buffered I/Q modulation receivercircuit, according to an embodiment of the present invention.

[0154]FIG. 100 illustrates an exemplary receiver with a placebo circuit,according to an embodiment of the present invention.

[0155]FIG. 101 shows an exemplary control signal waveform, and acorresponding exemplary placebo control signal waveform.

[0156]FIG. 102 illustrates a receiver with adj acent apertures circuit,according to an embodiment of the present invention.

[0157]FIG. 103 shows an exemplary control signal waveform, and acorresponding π-shifted control signal waveform.

[0158]FIG. 104 illustrates an exemplary receiver with adjacent aperturescircuit, according to an embodiment of the present invention.

[0159]FIG. 105 illustrates an exemplary circuit for improving dynamicrange, according to an embodiment of the present invention.

[0160] FIGS. 106A-C illustrate exemplary waveforms related to improvingdynamic range.

[0161]FIG. 107 illustrates an exemplary bias circuit, according to anembodiment of the present invention.

[0162]FIG. 108 depicts a flowchart that illustrates operational stepsfor down-converting and spectrally spreading an input signal, accordingto an embodiment of the present invention.

[0163]FIG. 109 depicts a flowchart that illustrates operational stepsfor down-converting an input signal and reducing a DC offset voltage,according to an embodiment of the present invention.

[0164]FIG. 110 depicts a flowchart that illustrates operational stepsfor reducing DC offset in a signal path, according to an embodiment ofthe present invention.

[0165]FIG. 111 depicts a flowchart that illustrates operational stepsfor down-converting an input signal and canceling DC offset voltages,according to an embodiment of the present invention.

[0166]FIG. 112 depicts a flowchart that illustrates operational stepsfor down-converting an input signal and canceling DC offset voltages,according to an embodiment of the present invention.

[0167]FIG. 13 depicts a flowchart that illustrates operational steps fordifferentially down-converting an input signal, according to anembodiment of the present invention.

[0168]FIG. 114 depicts a flowchart that illustrates operational stepsfor down-converting an input signal with a variety of control signalpulse widths, according to an embodiment of the present invention.

[0169]FIG. 115 depicts a flowchart that illustrates operational stepsfor down-converting an RF I/Q modulated input signal, according to anembodiment of the present invention.

[0170]FIG. 116 depicts a flowchart that illustrates operational stepsfor down-converting an RF I/Q modulated input signal, according to anembodiment of the present invention.

[0171]FIG. 117 depicts a flowchart that illustrates operational stepsfor down-converting an input signal and altering circuit re-radiation,according to an embodiment of the present invention.

[0172] FIGS. 118A-B depict flowcharts that illustrate operational stepsfor down-converting an input signal and altering circuit re-radiation,according to an embodiment of the present invention.

[0173]FIG. 119 depicts a flowchart that illustrates operational stepsfor improving dynamic range, according to an embodiment of the presentinvention.

[0174]FIG. 120 depicts a flowchart that illustrates operational stepsfor down-converting a RF I/Q modulated signal and reducing DC offsetvoltages, according to an embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0175] Table of Contents 1. Overview of the Invention 2. UniversalFrequency Translation 3. Frequency Down-conversion 3.1 Optional EnergyTransfer Signal Module 3.2 Smoothing the Down-Converted Signal 3.3Impedance Matching 3.4 Tanks and Resonant Structures 3.5 Charge andPower Transfer Concepts 3.6 Optimizing and Adjusting the Non-NegligibleAperture Width/Duration 3.6.1 Varying Input and Output Impedances 3.6.2Real Time Aperture Control 3.7 Adding a Bypass Network 3.8 Modifying theEnergy Transfer Signal Utilizing Feedback 3.9 Other Implementations 3.10Example Energy Transfer Down-Converters 4. Frequency Up-conversion 5.Enhanced Signal Reception 6. Unified Down-conversion and Filtering 7.Example Application Embodiments of the Invention 7.0 DC Offset,Re-radiation, and Dynamic Range Considerations and Corrections 7.1Overview of DC Offset and Re-radiation 7.1.1 Introduction 7.1.2 A BasicDC Offset Model 7.1.3 Clock Modulation via PN Code 7.1.3.1Interpretation of R_(xx)(T) and Required Leakage 7.1.3.2 Charge InjectedDC Offset 7.1.3.3 Clock Waveform Impact on CI Induced Offsets 7.1.3.4Bench Example 7.1.3.5 Complementary Architecture 7.1.3.6 Spreading CodeResults 7.1.4 UFD Module DC Offsets from Non-Linearities 7.2 ExampleEmbodiments to Address DC Offset and Re- radiation Problems 7.2.1 DCOffset 7.2.1.1 Reducing DC Offset by Spectral Spreading and De-spreading7.2.1.1.1 Conventional Wireless Communications Receiver 7.2.1.1.2Spread/De-spread Receiver Embodiment of the Present Invention 7.2.1.3Charge Injection Reduction Embodiment 7.2.1.4 Auto-Zero Compensation7.2.1.5 Reducing DC Offset with Differential Configurations 7.2.1.6Reducing DC Offset with Differential Outputs 7.2.2 Re-radiation 7.2.2.1Reducing Re-radiation by Adjusting Control Signal Attributes 7.2.2.1.1I/Q Modulation Receiver Control Signal Considerations and Embodiments7.2.2.1.1.1 Non-overlapping I/Q Control Signal Pulses Embodiments7.2.2.1.1.2 Buffered I/Q Modulation Receiver Embodiment 7.2.2.2 ReducingRe-radiation with Placebo Down-conversion Modules 7.2.2.3 ReducingRe-radiation with Adjacent Apertures 7.2.3 Additional DC Offset andRe-radiation Reduction Embodiments 7.3 Example Embodiments to ImproveDynamic Range 7.3.1 Adjusting Down-conversion Module Dynamic Range 7.4Example Receiver and Transmitter Embodiments for Addressing DC Offsetand Re-radiation 7.4.1 Example I/Q Modulation Receiver Embodiments7.4.1.1 Example I/Q Modulation Control Signal Generator Embodiments7.4.1.2 Detailed Example I/Q Modulation Receiver Embodiment withExemplary Waveforms 7.4.1.3 Example Single Channel Receiver Embodiment7.4.1.4 Alternative Example I/Q Modulation Receiver Embodiment 7.4.1.5Example Transmitter Embodiment 8. Conclusion 1. Overview of theInvention

[0176] The present invention is directed to receivers implemented usinguniversal frequency translation (UFT) modules. The UFT modules performfrequency translation operations. Embodiments of the present inventionincorporating various applications of the UFT module are describedbelow.

[0177] Receivers exhibit multiple advantages by using UFT modules. Theseadvantages include, but are not limited to, lower power consumption,longer power source life, fewer parts, lower required package size,lower package weight, lower cost, less tuning, and more effective signaltransmission and reception. The receivers of the present invention canreceive and transmit signals across a broad frequency range.Furthermore, the DC offset voltages and re-radiation generated byreceivers are the present invention are reduced or eliminated inembodiments. The structure and operation of embodiments of the UFTmodule, and various applications of the same, utilizing DCoffset/re-radiation reduction, are described in detail in the followingsections.

[0178] 2. Universal Frequency Translation

[0179] The present invention is related to frequency translation, andapplications of same. Such applications include, but are not limited to,frequency down-conversion, frequency up-conversion, enhanced signalreception, unified down-conversion and filtering, and combinations andapplications of same.

[0180]FIG. 1A illustrates a universal frequency translation (UFT) module102 according to embodiments of the invention. (The UFT module is alsosometimes called a universal frequency translator, or a universaltranslator.) As indicated by the example of FIG. 1A, some embodiments ofthe UFT module 102 include three ports (nodes), designated in FIG. 1A asPort 1, Port 2, and Port 3. Other UFT embodiments include other thanthree ports.

[0181] Generally, the UFT module 102 perhaps in combination with othercomponents) operates to generate an output signal from an input signal,where the frequency of the output signal differs from the frequency ofthe input signal. In other words, the UFT module 102 (and perhaps othercomponents) operates to generate the output signal from the input signalby translating the frequency (and perhaps other characteristics) of theinput signal to the frequency (and perhaps other characteristics) of theoutput signal.

[0182] An example embodiment of the UFT module 103 is generallyillustrated in FIG. 1B. Generally, the UFT module 103 includes a switch106 controlled by a control signal 108. The switch 106 is said to be acontrolled switch.

[0183] As noted above, some UFT embodiments include other than threeports. For example, and without limitation, FIG. 2 illustrates anexample UFT module 202. The example UFT module 202 includes a diode 204having two ports, designated as Port 1 and Port 2/3. This embodimentdoes not include a third port, as indicated by the dotted line aroundthe “Port 3” label.

[0184] The UFT module is a very powerful and flexible device. Itsflexibility is illustrated, in part, by the wide range of applicationsin which it can be used. Its power is illustrated, in part, by theusefulness and performance of such applications.

[0185] For example, a UFT module 115 can be used in a universalfrequency down-conversion (UFD) module 114, an example of which is shownin FIG. 1C. In this capacity, the UFT module 115 frequency down-convertsan input signal to an output signal.

[0186] As another example, as shown in FIG. 1D, a UFT module 117 can beused in a universal frequency up-conversion (UFU) module 116. In thiscapacity, the UFT module 117 frequency up-converts an input signal to anoutput signal.

[0187] These and other applications of the UFT module are describedbelow. Additional applications of the UFT module will be apparent topersons skilled in the relevant art(s) based on the teachings containedherein. In some applications, the UFT module is a required component. Inother applications, the UFT module is an optional component.

[0188] 3. Frequency Down-Conversion

[0189] The present invention is directed to systems and methods ofuniversal frequency down-conversion, and applications of same.

[0190] In particular, the following discussion describes down-convertingusing a Universal Frequency Translation Module. The down-conversion ofan EM signal by aliasing the EM signal at an aliasing rate is fullydescribed in co-pending U.S. Patent Application entitled “Method andSystem for Down-Converting Electromagnetic Signals,” Ser. No.09/176,022, filed Oct. 21, 1998, the full disclosure of which isincorporated herein by reference. A relevant portion of the abovementioned patent application is summarized below to describedown-converting an input signal to produce a down-converted signal thatexists at a lower frequency or a baseband signal.

[0191]FIG. 20A illustrates an aliasing module 2000 for down-conversionusing a universal frequency translation (UFT) module 2002 whichdown-converts an EM input signal 2004. In particular embodiments,aliasing module 2000 includes a switch 2008 and a capacitor 2010. Theelectronic alignment of the circuit components is flexible. That is, inone implementation, the switch 2008 is in series with input signal 2004and capacitor 2010 is shunted to ground (although it may be other thanground in configurations such as differential mode). In a secondimplementation (see FIG. 20G), the capacitor 2010 is in series with theinput signal 2004 and the switch 2008 is shunted to ground (although itmay be other than ground in configurations such as differential mode).Aliasing module 2000 with UFT module 2002 can be easily tailored todown-convert a wide variety of electromagnetic signals using aliasingfrequencies that are well below the frequencies of the EM input signal2004.

[0192] In one implementation, aliasing module 2000 down-converts theinput signal 2004 to an intermediate frequency (IF) signal. In anotherimplementation, the aliasing module 2000 down-converts the input signal2004 to a demodulated baseband signal. In yet another implementation,the input signal 2004 is a frequency modulated (FM) signal, and thealiasing module 2000 down-converts it to a non-FM signal, such as aphase modulated (PM) signal or an amplitude modulated (AM) signal. Eachof the above implementations is described below.

[0193] In an embodiment, the control signal 2006 includes a train ofpulses that repeat at an aliasing rate that is equal to, or less than,twice the frequency of the input signal 2004. In this embodiment, thecontrol signal 2006 is referred to herein as an aliasing signal becauseit is below the Nyquist rate for the frequency of the input signal 2004.Preferably, the frequency of control signal 2006 is much less than theinput signal 2004.

[0194] A train of pulses 2018 as shown in FIG. 20D controls the switch2008 to alias the input signal 2004 with the control signal 2006 togenerate a down-converted output signal 2012. More specifically, in anembodiment, switch 2008 closes on a first edge of each pulse 2020 ofFIG. 20D and opens on a second edge of each pulse. When the switch 2008is closed, the input signal 2004 is coupled to the capacitor 2010, andcharge is transferred from the input signal to the capacitor 2010. Thecharge stored during successive pulses forms down-converted outputsignal 2012.

[0195] Exemplary waveforms are shown in FIGS. 20B-20F.

[0196]FIG. 20B illustrates an analog amplitude modulated (AM) carriersignal 2014 that is an example of input signal 2004. For illustrativepurposes, in FIG. 20C, an analog AM carrier signal portion 2016illustrates a portion of the analog AM carrier signal 2014 on anexpanded time scale. The analog AM carrier signal portion 2016illustrates the analog AM carrier signal 2014 from time to t₀ time t₁.

[0197]FIG. 20D illustrates an exemplary aliasing signal 2018 that is anexample of control signal 2006. Aliasing signal 2018 is on approximatelythe same time scale as the analog AM carrier signal portion 2016. In theexample shown in FIG. 20D, the aliasing signal 2018 includes a train ofpulses 2020 having negligible apertures that tend towards zero (theinvention is not limited to this embodiment, as discussed below). Thepulse aperture may also be referred to as the pulse width as will beunderstood by those skilled in the art(s). The pulses 2020 repeat at analiasing rate, or pulse repetition rate of aliasing signal 2018. Thealiasing rate is determined as described below, and further described inco-pending U.S. Patent Application entitled “Method and System forDown-converting Electromagnetic Signals,” Ser. No. 09/176,022.

[0198] As noted above, the train of pulses 2020 (i.e., control signal2006) control the switch 2008 to alias the analog AM carrier signal 2016(i.e., input signal 2004) at the aliasing rate of the aliasing signal2018. Specifically, in this embodiment, the switch 2008 closes on afirst edge of each pulse and opens on a second edge of each pulse. Whenthe switch 2008 is closed, input signal 2004 is coupled to the capacitor2010, and charge is transferred from the input signal 2004 to thecapacitor 2010. The charge transferred during a pulse is referred toherein as an under-sample. Exemplary under-samples 2022 formdown-converted signal portion 2024 (FIG. 20E) that corresponds to theanalog AM carrier signal portion 2016 (FIG. 20C) and the train of pulses2020 (FIG. 20D). The charge stored during successive under-samples of AMcarrier signal 2014 form the down-converted signal 2024 (FIG. 20E) thatis an example of down-converted output signal 2012 (FIG. 20A). In FIG.20F, a demodulated baseband signal 2026 represents the demodulatedbaseband signal 2024 after filtering on a compressed time scale. Asillustrated, down-converted signal 2026 has substantially the same“amplitude envelope” as AM carrier signal 2014. Therefore, FIGS. 20B-20Fillustrate down-conversion of AM carrier signal 2014.

[0199] The waveforms shown in FIGS. 20B-20F are discussed herein forillustrative purposes only, and are not limiting. Additional exemplarytime domain and frequency domain drawings, and exemplary methods andsystems of the invention relating thereto, are disclosed in co-pendingU.S. Patent Application entitled “Method and System for Down-convertingElectromagnetic Signals,” Ser. No. 09/176,022.

[0200] The aliasing rate of control signal 2006 determines whether theinput signal 2004 is down-converted to an IF signal, down-converted to ademodulated baseband signal, or down-converted from an FM signal to a PMor an AM signal. Generally, relationships between the input signal 2004,the aliasing rate of the control signal 2006, and the down-convertedoutput signal 2012 are illustrated below:

(Freq. of input signal 2004)=n·(Freq. of control signal 2006)±(Freq. ofdown-converted output signal 2012)

[0201] For the examples contained herein, only the “+” condition will bediscussed. The value of n represents a harmonic or sub-harmonic of inputsignal 2004 (e.g., n=0.5, 1, 2, 3, . . . ).

[0202] When the aliasing rate of control signal 2006 is off-set from thefrequency of input signal 2004, or off-set from a harmonic orsub-harmonic thereof, input signal 2004 is down-converted to an IFsignal. This is because the under-sampling pulses occur at differentphases of subsequent cycles of input signal 2004. As a result, theunder-samples form a lower frequency oscillating pattern. If the inputsignal 2004 includes lower frequency changes, such as amplitude,frequency, phase, etc., or any combination thereof, the charge storedduring associated under-samples reflects the lower frequency changes,resulting in similar changes on the down-converted IF signal. Forexample, to down-convert a 901 MHZ input signal to a 1 MHZ IF signal,the frequency of the control signal 2006 would be calculated as follows:

(Freq _(input) −Freq _(IF))/n=Freq _(control)

(901 MHZ−1 MHZ)/n=900/n

[0203] For n=0.5, 1, 2, 3, 4, etc., the frequency of the control signal2006 would be substantially equal to 1.8 GHz, 900 MHZ, 450 MHZ, 300 MHZ,225 MHZ, etc.

[0204] Exemplary time domain and frequency domain drawings, illustratingdown-conversion of analog and digital AM, PM and FM signals to IFsignals, and exemplary methods and systems thereof, are disclosed inco-pending U.S. Patent Application entitled “Method and System forDown-converting Electromagnetic Signals,” Ser. No. 09/176,022.

[0205] Alternatively, when the aliasing rate of the control signal 2006is substantially equal to the frequency of the input signal 2004, orsubstantially equal to a harmonic or sub-harmonic thereof, input signal2004 is directly down-converted to a demodulated baseband signal. Thisis because, without modulation, the under-sampling pulses occur at thesame point of subsequent cycles of the input signal 2004. As a result,the under-samples form a constant output baseband signal. If the inputsignal 2004 includes lower frequency changes, such as amplitude,frequency, phase, etc., or any combination thereof, the charge storedduring associated under-samples reflects the lower frequency changes,resulting in similar changes on the demodulated baseband signal. Forexample, to directly down-convert a 900 MHZ input signal to ademodulated baseband signal (i.e., zero IF), the frequency of thecontrol signal 2006 would be calculated as follows:

(Freq _(input) −Freq _(IF))/n=Freq _(control)

(900 MHZ−0 MHZ)/n=900 MHZ/n

[0206] For n=0.5, 1, 2, 3, 4, etc., the frequency of the control signal2006 should be substantially equal to 1.8 GHz, 900 MHZ, 450 MHZ, 300MHZ, 225 MHZ, etc.

[0207] Exemplary time domain and frequency domain drawings, illustratingdirect down-conversion of analog and digital AM and PM signals todemodulated baseband signals, and exemplary methods and systems thereof,are disclosed in the co-pending U.S. Patent Application entitled “Methodand System for Down-converting Electromagnetic Signals,” Ser. No.09/176,022.

[0208] Alternatively, to down-convert an input FM signal to a non-FMsignal, a frequency within the FM bandwidth must be down-converted tobaseband (i.e., zero IF). As an example, to down-convert a frequencyshift keying (FSK) signal (a sub-set of FM) to a phase shift keying(PSK) signal (a subset of PM), the mid-point between a lower frequencyF₁ and an upper frequency F₂ (that is, [(F₁+F₂)÷2]) of the FSK signal isdown-converted to zero IF. For example, to down-convert an FSK signalhaving F₁ equal to 899 MHZ and F₂ equal to 901 MHZ, to a PSK signal, thealiasing rate of the control signal 2006 would be calculated as follows:

Frequency of the input=(F ₁ +F ₂)÷2=(899 MHZ+901 MHZ)÷2=900 MHZ

[0209] Frequency of the down-converted signal=0 (i.e., baseband)

(Freq _(input) −Freq _(IF))/n=Freq _(control)

(900 MHZ−0 MHZ)/n=900 MHZ/n

[0210] For n=0.5, 1, 2, 3, etc., the frequency of the control signal2006 should be substantially equal to 1.8 GHz, 900 MHZ, 450 MHZ, 300MHZ, 225 MHZ, etc. The frequency of the down-converted PSK signal issubstantially equal to one half the difference between the lowerfrequency F₁ and the upper frequency F₂.

[0211] As another example, to down-convert a FSK signal to an amplitudeshift keying (ASK) signal (a subset of AM), either the lower frequencyF₁ or the upper frequency F₂ of the FSK signal is down-converted to zeroIF. For example, to down-convert an FSK signal having F₁ equal to 900MHZ and F₂ equal to 901 MHZ, to an ASK signal, the aliasing rate of thecontrol signal 2006 should be substantially equal to:

(900 MHZ−0 MHZ)/n=900 MHZ/n, or

(901 MHZ−0 MHZ)/n=901 MHZ/n.

[0212] For the former case of 900 MHZ/n, and for n=0.5, 1, 2, 3, 4,etc., the frequency of the control signal 2006 should be substantiallyequal to 1.8 GHz, 900 MHZ, 450 MHZ, 300 MHZ, 225 MHZ, etc. For thelatter case of 901 MHZ/n, and for n=0.5, 1, 2, 3,4, etc., the frequencyof the control signal 2006 should be substantially equal to 1.802 GHz,901 MHZ, 450.5 MHZ, 300.333 MHZ, 225.25 MHZ, etc. The frequency of thedown-converted AM signal is substantially equal to the differencebetween the lower frequency F₁ and the upper frequency F₂ (i.e., 1 MHZ).

[0213] Exemplary time domain and frequency domain drawings, illustratingdown-conversion of FM signals to non-FM signals, and exemplary methodsand systems thereof, are disclosed in the co-pending U.S. PatentApplication entitled “Method and System for Down-convertingElectromagnetic Signals,” Ser. No. 09/176,022.

[0214] In an embodiment, the pulses of the control signal 2006 havenegligible apertures that tend towards zero. This makes the UFT module2002 a high input impedance device. This configuration is useful forsituations where minimal disturbance of the input signal may be desired.

[0215] In another embodiment, the pulses of the control signal 2006 havenon-negligible apertures that tend away from zero. This makes the UFTmodule 2002 a lower input impedance device. This allows the lower inputimpedance of the UFT module 2002 to be substantially matched with asource impedance of the input signal 2004. This also improves the energytransfer from the input signal 2004 to the down-converted output signal2012, and hence the efficiency and signal to noise (s/n) ratio of UFTmodule 2002.

[0216] Exemplary systems and methods for generating and optimizing thecontrol signal 2006 and for otherwise improving energy transfer and s/nratio, are disclosed in the co-pending U.S. Patent Application entitled“Method and System for Down-converting Electromagnetic Signals,” Ser.No. 09/176,022.

[0217] When the pulses of the control signal 2006 have non-negligibleapertures, the aliasing module 2000 is referred to interchangeablyherein as an energy transfer module or a gated transfer module, and thecontrol signal 2006 is referred to as an energy transfer signal.Exemplary systems and methods for generating and optimizing the controlsignal 2006 and for otherwise improving energy transfer and/or signal tonoise ratio in an energy transfer module are described below.

[0218] 3.1. Optional Energy Transfer Signal Module

[0219]FIG. 47 illustrates an energy transfer system 4701 that includesan optional energy transfer signal module 4702, which can perform any ofa variety of functions or combinations of functions including, but notlimited to, generating the energy transfer signal 4506.

[0220] In an embodiment, the optional energy transfer signal module 4702includes an aperture generator, an example of which is illustrated inFIG. 46C as an aperture generator 4620. The aperture generator 4620generates non-negligible aperture pulses 4626 from an input signal 4624.The input signal 4624 can be any type of periodic signal, including, butnot limited to, a sinusoid, a square wave, a saw-tooth wave, etc.Systems for generating the input signal 4624 are described below.

[0221] The width or aperture of the pulses 4626 is determined by delaythrough the branch 4622 of the aperture generator 4620. Generally, asthe desired pulse width increases, the difficulty in meeting therequirements of the aperture generator 4620 decrease. In other words, togenerate non-negligible aperture pulses for a given EM input frequency,the components utilized in the example aperture generator 4620 do notrequire as fast reaction times as those that are required in anunder-sampling system operating with the same EM input frequency.

[0222] The example logic and implementation shown in the aperturegenerator 4620 are provided for illustrative purposes only, and are notlimiting. The actual logic employed can take many forms. The exampleaperture generator 4620 includes an optional inverter 4628, which isshown for polarity consistency with other examples provided herein.

[0223] An example implementation of the aperture generator 4620 isillustrated in FIG. 46D. Additional examples of aperture generationlogic are provided in FIGS. 46A and 46B. FIG. 46A illustrates a risingedge pulse generator 4640, which generates pulses 4626 on rising edgesof the input signal 4624. FIG. 46B illustrates a falling edge pulsegenerator 4650, which generates pulses 4626 on falling edges of theinput signal 4624.

[0224] In an embodiment, the input signal 4624 is generated externallyof the energy transfer signal module 4702, as illustrated in FIG. 47.Alternatively, the input signal 4724 is generated internally by theenergy transfer signal module 4702. The input signal 4624 can begenerated by an oscillator, as illustrated in FIG. 46E by an oscillator4630. The oscillator 4630 can be internal to the energy transfer signalmodule 4702 or external to the energy transfer signal module 4702. Theoscillator 4630 can be external to the energy transfer system 4701. Theoutput of the oscillator 4630 may be any periodic waveform.

[0225] The type of down-conversion performed by the energy transfersystem 4701 depends upon the aliasing rate of the energy transfer signal4506, which is determined by the frequency of the pulses 4626. Thefrequency of the pulses 4626 is determined by the frequency of the inputsignal 4624. For example, when the frequency of the input signal 4624 issubstantially equal to a harmonic or a sub-harmonic of the EM signal4504, the EM signal 4504 is directly down-converted to baseband (e.g.when the EM signal is an AM signal or a PM signal), or converted from FMto a non-FM signal. When the frequency of the input signal 4624 issubstantially equal to a harmonic or a sub-harmonic of a differencefrequency, the EM signal 4504 is down-converted to an intermediatesignal.

[0226] The optional energy transfer signal module 4702 can beimplemented in hardware, software, firmware, or any combination thereof.

[0227] 3.2 Smoothing the Down-Converted Signal

[0228] Referring back to FIG. 20A, the down-converted output signal 2012may be smoothed by filtering as desired.

[0229] 3.3. Impedance Matching

[0230] The energy transfer module 2000 has input and output impedancesgenerally defined by (1) the duty cycle of the switch module (i.e., UFT2002), and (2) the impedance of the storage module (e.g., capacitor2010), at the frequencies of interest (e.g. at the EM input, andintermediate/baseband frequencies).

[0231] Starting with an aperture width of approximately ½ the period ofthe EM signal being down-converted as a preferred embodiment, thisaperture width (e.g. the “closed time”) can be decreased. As theaperture width is decreased, the characteristic impedance at the inputand the output of the energy transfer module increases. Alternatively,as the aperture width increases from ½ the period of the EM signal beingdown-converted, the impedance of the energy transfer module decreases.

[0232] One of the steps in determining the characteristic inputimpedance of the energy transfer module could be to measure its value.In an embodiment, the energy transfer module's characteristic inputimpedance is 300 ohms. An impedance matching circuit can be utilized toefficiently couple an input EM signal that has a source impedance of,for example, 50 ohms, with the energy transfer module's impedance of,for example, 300 ohms. Matching these impedances can be accomplished invarious manners, including providing the necessary impedance directly orthe use of an impedance match circuit as described below. Referring toFIG. 48, a specific embodiment using an RF signal as an input, assumingthat the impedance 4812 is a relatively low impedance of approximately50 Ohms, for example, and the input impedance 4816 is approximately 300Ohms, an initial configuration for the input impedance match module 4806can include an inductor 5006 and a capacitor 5008, configured as shownin FIG. 50. The configuration of the inductor 5006 and the capacitor5008 is a possible configuration when going from a low impedance to ahigh impedance. Inductor 5006 and the capacitor 5008 constitute an Lmatch, the calculation of the values which is well known to thoseskilled in the relevant arts.

[0233] The output characteristic impedance can be impedance matched totake into consideration the desired output frequencies. One of the stepsin determining the characteristic output impedance of the energytransfer module could be to measure its value. Balancing the very lowimpedance of the storage module at the input EM frequency, the storagemodule should have an impedance at the desired output frequencies thatis preferably greater than or equal to the load that is intended to bedriven (for example, in an embodiment, storage module impedance at adesired 1 MHz output frequency is 2K ohm and the desired load to bedriven is 50 ohms). An additional benefit of impedance matching is thatfiltering of unwanted signals can also be accomplished with the samecomponents.

[0234] In an embodiment, the energy transfer module's characteristicoutput impedance is 2K ohms. An impedance matching circuit can beutilized to efficiently couple the down-converted signal with an outputimpedance of, for example, 2K ohms, to a load of, for example, 50 ohms.Matching these impedances can be accomplished in various manners,including providing the necessary load impedance directly or the use ofan impedance match circuit as described below.

[0235] When matching from a high impedance to a low impedance, acapacitor 5014 and an inductor 5016 can be configured as shown in FIG.50. The capacitor 5014 and the inductor 5016 constitute an L match, thecalculation of the component values being well known to those skilled inthe relevant arts.

[0236] The configuration of the input impedance match module 4806 andthe output impedance match module 4808 are considered to be initialstarting points for impedance matching, in accordance with the presentinvention. In some situations, the initial designs may be suitablewithout further optimization. In other situations, the initial designscan be optimized in accordance with other various design criteria andconsiderations.

[0237] As other optional optimizing structures and/or components areutilized, their affect on the characteristic impedance of the energytransfer module should be taken into account in the match along withtheir own original criteria.

[0238] 3.4 Tanks and Resonant Structures

[0239] Resonant tank and other resonant structures can be used tofurther optimize the energy transfer characteristics of the invention.For example, resonant structures, resonant about the input frequency,can be used to store energy from the input signal when the switch isopen, a period during which one may conclude that the architecture wouldotherwise be limited in its maximum possible efficiency. Resonant tankand other resonant structures can include, but are not limited to,surface acoustic wave (SAW) filters, dielectric resonators, diplexers,capacitors, inductors, etc.

[0240] An example embodiment is shown in FIG. 60A. Two additionalembodiments are shown in FIG. 55 and FIG. 63. Alternate implementationswill be apparent to persons skilled in the relevant art(s) based on theteachings contained herein. Alternate implementations fall within thescope and spirit of the present invention. These implementations takeadvantage of properties of series and parallel (tank) resonant circuits.

[0241]FIG. 60A illustrates parallel tank circuits in a differentialimplementation. A first parallel resonant or tank circuit consists of acapacitor 6038 and an inductor 6020 (tank1). A second tank circuitconsists of a capacitor 6034 and an inductor 6036 (tank2).

[0242] As is apparent to one skilled in the relevant art(s), paralleltank circuits provide:

[0243] low impedance to frequencies below resonance;

[0244] low impedance to frequencies above resonance; and

[0245] high impedance to frequencies at and near resonance.

[0246] In the illustrated example of FIG. 60A, the first and second tankcircuits resonate at approximately 920 MHz. At and near resonance, theimpedance of these circuits is relatively high. Therefore, in thecircuit configuration shown in FIG. 60A, both tank circuits appear asrelatively high impedance to the input frequency of 950 MHz, whilesimultaneously appearing as relatively low impedance to frequencies inthe desired output range of 50 MHz.

[0247] An energy transfer signal 6042 controls a switch 6014. When theenergy transfer signal 6042 controls the switch 6014 to open and close,high frequency signal components are not allowed to pass through tank1or tank2. However, the lower signal components (50 Mhz in thisembodiment) generated by the system are allowed to pass through tank1and tank2 with little attenuation. The effect of tank1 and tank2 is tofurther separate the input and output signals from the same node therebyproducing a more stable input and output impedance. Capacitors 6018 and6040 act to store the 50 MHz output signal energy between energytransfer pulses.

[0248] Further energy transfer optimization is provided by placing aninductor 6010 in series with a storage capacitor 6012 as shown. In theillustrated example, the series resonant frequency of this circuitarrangement is approximately 1 GHz. This circuit increases the energytransfer characteristic of the system. The ratio of the impedance ofinductor 6010 and the impedance of the storage capacitor 6012 ispreferably kept relatively small so that the majority of the energyavailable will be transferred to storage capacitor 6012 duringoperation. Exemplary output signals A and B are illustrated in FIGS. 60Band 60C, respectively.

[0249] In FIG. 60A, circuit components 6004 and 6006 form an inputimpedance match. Circuit components 6032 and 6030 form an outputimpedance match into a 50 ohm resistor 6028. Circuit components 6022 and6024 form a second output impedance match into a 50 ohm resistor 6026.Capacitors 6008 and 6012 act as storage capacitors for the embodiment.Voltage source 6046 and resistor 6002 generate a 950 MHz signal with a50 ohm output impedance, which are used as the input to the circuit.Circuit element 6016 includes a 150 MHz oscillator and a pulsegenerator, which are used to generate the energy transfer signal 6042.

[0250]FIG. 55 illustrates a shunt tank circuit 5510 in a single-endedto-single-ended system 5512. Similarly, FIG. 63 illustrates a shunt tankcircuit 6310 in a system 6312. The tank circuits 5510 and 6310 lowerdriving source impedance, which improves transient response. The tankcircuits 5510 and 6310 are able store the energy from the input signaland provide a low driving source impedance to transfer that energythroughout the aperture of the closed switch. The transient nature ofthe switch aperture can be viewed as having a response that, in additionto including the input frequency, has large component frequencies abovethe input frequency, (i.e. higher frequencies than the input frequencyare also able to effectively pass through the aperture). Resonantcircuits or structures, for example resonant tanks 5510 or 6310, cantake advantage of this by being able to transfer energy throughout theswitch's transient frequency response (i.e. the capacitor in theresonant tank appears as a low driving source impedance during thetransient period of the aperture).

[0251] The example tank and resonant structures described above are forillustrative purposes and are not limiting. Alternate configurations canbe utilized. The various resonant tanks and structures discussed can becombined or utilized independently as is now apparent.

[0252] 3.5 Charge and Power Transfer Concepts

[0253] Concepts of charge transfer are now described with reference toFIGS. 71A-F. FIG. 71A illustrates a circuit 7102, including a switch Sand a capacitor 7106 having a capacitance C. The switch S is controlledby a control signal 7108, which includes pulses 19010 having aperturesT.

[0254] In FIG. 71B, Equation 10 illustrates that the charge q on acapacitor having a capacitance C, such as the capacitor 7106, isproportional to the voltage V across the capacitor, where:

[0255] q=Charge in Coulombs

[0256] C=Capacitance in Farads

[0257] V=Voltage in Volts

[0258] A=Input Signal Amplitude

[0259] Where the voltage V is represented by Equation 11, Equation 10can be rewritten as Equation 12. The change in charge Δq over time t isillustrated as in Equation 13 as Δq(t), which can be rewritten asEquation 14. Using the sum-to-product trigonometric identity of Equation15, Equation 14 can be rewritten as Equation 16, which can be rewrittenas equation 17.

[0260] Note that the sin term in Equation 11 is a function of theaperture T only. Thus, Δq(t) is at a maximum when T is equal to an oddmultiple of π (i.e., π, 3π, 5π, . . . ). Therefore, the capacitor 7106experiences the greatest change in charge when the aperture T has avalue of π or a time interval representative of 180 degrees of the inputsinusoid. Conversely, when T is equal to 2π, 4π, . . . , minimal chargeis transferred.

[0261] Equations 18, 19, and 20 solve for q(t) by integrating Equation10, allowing the charge on the capacitor 7106 with respect to time to begraphed on the same axis as the input sinusoid sin(t), as illustrated inthe graph of FIG. 71C. As the aperture T decreases in value or tendstoward an impulse, the phase between the charge on the capacitor C orq(t) and sin(t) tend toward zero. This is illustrated in the graph ofFIG. 71D, which indicates that the maximum impulse charge transferoccurs near the input voltage maxima. As this graph indicates,considerably less charge is transferred as the value of T decreases.

[0262] Power/charge relationships are illustrated in Equations 21-26 ofFIG. 71E, where it is shown that power is proportional to charge, andtransferred charge is inversely proportional to insertion loss.

[0263] Concepts of insertion loss are illustrated in FIG. 71F.Generally, the noise figure of a lossy passive device is numericallyequal to the device insertion loss. Alternatively, the noise figure forany device cannot be less that its insertion loss. Insertion loss can beexpressed by Equation 27 or 28. From the above discussion, it isobserved that as the aperture T increases, more charge is transferredfrom the input to the capacitor 7106, which increases power transferfrom the input to the output. It has been observed that it is notnecessary to accurately reproduce the input voltage at the outputbecause relative modulated amplitude and phase information is retainedin the transferred power.

[0264] 3.6 Optimizing and Adjusting the Non-Negligible ApertureWidth/Duration

[0265] 3.6.1 Varying Input and Output Impedances

[0266] In an embodiment of the invention, the energy transfer signal(i.e., control signal 2006 in FIG. 20A), is used to vary the inputimpedance seen by the EM Signal 2004 and to vary the output impedancedriving a load. An example of this embodiment is described below using agated transfer module 5101 shown in FIG. 51A. The method described belowis not limited to the gated transfer module 5101.

[0267] In FIG. 51A, when switch 5106 is closed, the impedance lookinginto circuit 5102 is substantially the impedance of a storage module,illustrated here as a storage capacitance 5108, in parallel with theimpedance of a load 5112. When the switch 5106 is open, the impedance atpoint 5114 approaches infinity. It follows that the average impedance atpoint 5114 can be varied from the impedance of the storage moduleillustrated in parallel with the load 5112, to the highest obtainableimpedance when switch 5106 is open, by varying the ratio of the timethat switch 5106 is open to the time switch 5106 is closed. The switch5106 is controlled by an energy transfer signal 5110. Thus the impedanceat point 5114 can be varied by controlling the aperture width of theenergy transfer signal in conjunction with the aliasing rate.

[0268] An example method of altering the energy transfer signal 5106 ofFIG. 51A is now described with reference to FIG. 49A, where a circuit4902 receives an input oscillating signal 4906 and outputs a pulse trainshown as doubler output signal 4904. The circuit 4902 can be used togenerate the energy transfer signal 5106. Example waveforms of 4904 areshown on FIG. 49C.

[0269] It can be shown that by varying the delay of the signalpropagated by the inverter 4908, the width of the pulses in the doubleroutput signal 4904 can be varied. Increasing the delay of the signalpropagated by inverter 4908, increases the width of the pulses. Thesignal propagated by inverter 4908 can be delayed by introducing a R/Clow pass network in the output of inverter 4908. Other means of alteringthe delay of the signal propagated by inverter 4908 will be well knownto those skilled in the art.

[0270] 3.6.2 Real Time Aperture Control

[0271] In an embodiment, the aperture width/duration is adjusted in realtime. For example, referring to the timing diagrams in FIGS. 64B-F, aclock signal 6414 (FIG. 64B) is utilized to generate an energy transfersignal 6416 (FIG. 64F), which includes energy transfer pluses 6418,having variable apertures 6420. In an embodiment, the clock signal 6414is inverted as illustrated by inverted clock signal 6422 (FIG. 64D). Theclock signal 6414 is also delayed, as illustrated by delayed clocksignal 6424 (FIG. 64E). The inverted clock signal 6414 and the delayedclock signal 6424 are then ANDed together, generating an energy transfersignal 6416, which is active—energy transfer pulses 6418—when thedelayed clock signal 6424 and the inverted clock signal 6422 are bothactive. The amount of delay imparted to the delayed clock signal 6424substantially determines the width or duration of the apertures 6420. Byvarying the delay in real time, the apertures are adjusted in real time.

[0272] In an alternative implementation, the inverted clock signal 6422is delayed relative to the original clock signal 6414, and then ANDedwith the original clock signal 6414. Alternatively, the original clocksignal 6414 is delayed then inverted, and the result ANDed with theoriginal clock signal 6414.

[0273]FIG. 64A illustrates an exemplary real time aperture controlsystem 6402 that can be utilized to adjust apertures in real time. Theexample real time aperture control system 6402 includes an RC circuit6404, which includes a voltage variable capacitor 6412 and a resistor6426. The real time aperture control system 6402 also includes aninverter 6406 and an AND gate 6408. The AND gate 6408 optionallyincludes an enable input 6410 for enabling/disabling the AND gate 6408.The RC circuit 6404. The real time aperture control system 6402optionally includes an amplifier 6428.

[0274] Operation of the real time aperture control circuit is describedwith reference to the timing diagrams of FIGS. 64B-F. The real timecontrol system 6402 receives the input clock signal 6414, which isprovided to both the inverter 6406 and to the RC circuit 6404. Theinverter 6406 outputs the inverted clock signal 6422 and presents it tothe AND gate 6408. The RC circuit 6404 delays the clock signal 6414 andoutputs the delayed clock signal 6424. The delay is determined primarilyby the capacitance of the voltage variable capacitor 6412. Generally, asthe capacitance decreases, the delay decreases.

[0275] The delayed clock signal 6424 is optionally amplified by theoptional amplifier 6428, before being presented to the AND gate 6408.Amplification is desired, for example, where the RC constant of the RCcircuit 6404 attenuates the signal below the threshold of the AND gate6408.

[0276] The AND gate 6408 ANDs the delayed clock signal 6424, theinverted clock signal 6422, and the optional Enable signal 6410, togenerate the energy transfer signal 6416. The apertures 6420 areadjusted in real time by varying the voltage to the voltage variablecapacitor 6412.

[0277] In an embodiment, the apertures 6420 are controlled to optimizepower transfer. For example, in an embodiment, the apertures 6420 arecontrolled to maximize power transfer. Alternatively, the apertures 6420are controlled for variable gain control (e.g. automatic gain control -AGC). Inthis embodiment, power transfer is reduced by reducing theapertures 6420.

[0278] As can now be readily seen from this disclosure, many of theaperture circuits presented, and others, can be modified as in circuitsillustrated in FIGS. 46H-K. Modification or selection of the aperturecan be done at the design level to remain a fixed value in the circuit,or in an alternative embodiment, may be dynamically adjusted tocompensate for, or address, various design goals such as receiving RFsignals with enhanced efficiency that are in distinctively differentbands o foperation, e.g. RF signals at 900 MHZ and 1.8 GHz.

[0279] 3.7 Adding a Bypass Network

[0280] In an embodiment of the invention, a bypass network is added toimprove the efficiency of the energy transfer module. Such a bypassnetwork can be viewed as a means of synthetic aperture widening.Components for a bypass network are selected so that the bypass networkappears substantially lower impedance to transients of the switch module(i.e., frequencies greater than the received EM signal) and appears as amoderate to high impedance to the input EM signal (e.g., greater that100 Ohms at the RF frequency).

[0281] The time that the input signal is now connected to the oppositeside of the switch module is lengthened due to the shaping caused bythis network, which in simple realizations may be a capacitor or seriesresonant inductor-capacitor. A network that is series resonant above theinput frequency would be a typical implementation. This shaping improvesthe conversion efficiency of an input signal that would otherwise, ifone considered the aperture of the energy transfer signal only, berelatively low in frequency to be optimal.

[0282] For example, referring to FIG. 61 a bypass network 6102 (shown inthis instance as capacitor 6112), is shown bypassing switch module 6104.In this embodiment the bypass network increases the efficiency of theenergy transfer module when, for example, less than optimal aperturewidths were chosen for a given input frequency on the energy transfersignal 6106. The bypass network 6102 could be of differentconfigurations than shown in FIG. 61. Such an alternate is illustratedin FIG. 57. Similarly, FIG. 62 illustrates another example bypassnetwork 6202, including a capacitor 6204.

[0283] The following discussion will demonstrate the effects of aminimized aperture and the benefit provided by a bypassing network.Beginning with an initial circuit having a 550 ps aperture in FIG. 65,its output is seen to be 2.8 mVpp applied to a 50 ohm load in FIG. 69A.Changing the aperture to 270 ps as shown in FIG. 66 results in adiminished output of 2.5 Vpp applied to a 50 ohm load as shown in FIG.69B. To compensate for this loss, a bypass network may be added, aspecific implementation is provided in FIG. 67. The result of thisaddition is that 3.2 Vpp can now be applied to the 50 ohm load as shownin FIG. 70A. The circuit with the bypass network in FIG. 67 also hadthree values adjusted in the surrounding circuit to compensate for theimpedance changes introduced by the bypass network and narrowedaperture. FIG. 68 verifies that those changes added to the circuit, butwithout the bypass network, did not themselves bring about the increasedefficiency demonstrated by the embodiment in FIG. 67 with the bypassnetwork. FIG. 70B shows the result of using the circuit in FIG. 68 inwhich only 1.88 Vpp was able to be applied to a 50 ohm load.

[0284] 3.8 Modifying the Energy Transfer Signal Utilizing Feedback

[0285]FIG. 47 shows an embodiment of a system 4701 which usesdown-converted Signal 4708B as feedback 4706 to control variouscharacteristics of the energy transfer module 4704 to modify thedown-converted signal 4708B.

[0286] Generally, the amplitude of the down-converted signal 4708Bvaries as a function of the frequency and phase differences between theEM signal 4504 and the energy transfer signal 4506. In an embodiment,the down-converted signal 4708B is used as the feedback 4706 to controlthe frequency and phase relationship between the EM signal 4504 and theenergy transfer signal 4506. This can be accomplished using the examplelogic in FIG. 52A. The example circuit in FIG. 52A can be included inthe energy transfer signal module 4702. Alternate implementations willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. Alternate implementations fall within thescope and spirit of the present invention. In this embodiment astate-machine is used as an example.

[0287] In the example of FIG. 52A, a state machine 5204 reads an analogto digital converter, A/D 5202, and controls a digital to analogconverter, DAC 5206. In an embodiment, the state machine 5204 includes 2memory locations, Previous and Current, to store and recall the resultsof reading A/D 5202. In an embodiment, the state machine 5204 utilizesat least one memory flag.

[0288] The DAC 5206 controls an input to a voltage controlledoscillator, VCO 5208. VCO 5208 controls a frequency input of a pulsegenerator 5210, which, in an embodiment, is substantially similar to thepulse generator shown in FIG. 46C. The pulse generator 5210 generatesenergy transfer signal 4506.

[0289] In an embodiment, the state machine 5204 operates in accordancewith a state machine flowchart 5219 in FIG. 52B. The result of thisoperation is to modify the frequency and phase relationship between theenergy transfer signal 4506 and the EM signal 4504, to substantiallymaintain the amplitude of the down-converted signal 4708B at an optimumlevel.

[0290] The amplitude of the down-converted signal 4708B can be made tovary with the amplitude of the energy transfer signal 4506. In anembodiment where the switch module 6502 is a FET as shown in FIG. 45A,wherein the gate 4518 receives the energy transfer signal 4506, theamplitude of the energy transfer signal 4506 can determine the “on”resistance of the FET, which affects the amplitude of the down-convertedsignal 4708B. The energy transfer signal module 4702, as shown in FIG.52C, can be an analog circuit that enables an automatic gain controlfunction. Alternate implementations will be apparent to persons skilledin the relevant art(s) based on the teachings contained herein.Alternate implementations fall within the scope and spirit of thepresent invention.

[0291] 3.9 Other Implementations

[0292] The implementations described above are provided for purposes ofillustration. These implementations are not intended to limit theinvention. Alternate implementations, differing slightly orsubstantially from those described herein, will be apparent to personsskilled in the relevant art(s) based on the teachings contained herein.Such alternate implementations fall within the scope and spirit of thepresent invention.

[0293] 3.10 Example Energy Transfer Down-Converters

[0294] Example implementations are described below for illustrativepurposes. The invention is not limited to these examples.

[0295]FIG. 53 is a schematic diagram of an exemplary circuit to downconvert a 915 MHZ signal to a 5 MHZ signal using a 101.1 MHZ clock.

[0296]FIG. 54 shows example simulation waveforms for the circuit of FIG.53. Waveform 5302 is the input to the circuit showing the distortionscaused by the switch closure. Waveform 5304 is the unfiltered output atthe storage unit. Waveform 5306 is the impedance matched output of thedown-converter on a different time scale.

[0297]FIG. 55 is a schematic diagram of an exemplary circuit todown-convert a 915 MHZ signal to a 5 MHZ signal using a 101.1 MHZ clock.The circuit has additional tank circuitry to improve conversionefficiency.

[0298]FIG. 56 shows example simulation waveforms for the circuit of FIG.55. Waveform 5502 is the input to the circuit showing the distortionscaused by the switch closure. Waveform 5504 is the unfiltered output atthe storage unit. Waveform 5506 is the output of the down-converterafter the impedance match circuit.

[0299]FIG. 57 is a schematic diagram of an exemplary circuit todown-convert a 915 MHZ signal to a 5 MHZ signal using a 101.1 MHZ clock.The circuit has switch bypass circuitry to improve conversionefficiency.

[0300]FIG. 58 shows example simulation waveforms for the circuit of FIG.57. Waveform 5702 is the input to the circuit showing the distortionscaused by the switch closure. Waveform 5704 is the unfiltered output atthe storage unit. Waveform 5706 is the output of the down-converterafter the impedance match circuit.

[0301]FIG. 59 shows a schematic of the example circuit in FIG. 53connected to an FSK source that alternates between 913 and 917 MHZ, at abaud rate of 500 Kbaud. FIG. 72 shows the original FSK waveform 5902 andthe down-converted waveform 5904 at the output of the load impedancematch circuit.

[0302] 4. Frequency Up-Conversion

[0303] The present invention is directed to systems and methods offrequency up-conversion, and applications of same.

[0304] An example frequency up-conversion system 300 is illustrated inFIG. 3. The frequency up-conversion system 300 is now described.

[0305] An input signal 302 (designated as “Control Signal” in FIG. 3) isaccepted by a switch module 304. For purposes of example only, assumethat the input signal 302 is a FM input signal 606, an example of whichis shown in FIG. 6C. FM input signal 606 may have been generated bymodulating information signal 602 onto oscillating signal 604 (FIGS. 6Aand 6B). It should be understood that the invention is not limited tothis embodiment. The information signal 602 can be analog, digital, orany combination thereof, and any modulation scheme can be used.

[0306] The output of switch module 304 is a harmonically rich signal306, shown for example in FIG. 6D as a harmonically rich signal 608. Theharmonically rich signal 608 has a continuous and periodic waveform.

[0307]FIG. 6E is an expanded view of two sections of harmonically richsignal 608, section 610 and section 612. The harmonically rich signal608 may be a rectangular wave, such as a square wave or a pulse(although, the invention is not limited to this embodiment). For ease ofdiscussion, the term “rectangular waveform” is used to refer towaveforms that are substantially rectangular. In a similar manner, theterm “square wave” refers to those waveforms that are substantiallysquare and it is not the intent of the present invention that a perfectsquare wave be generated or needed.

[0308] Harmonically rich signal 608 is comprised of a plurality ofsinusoidal waves whose frequencies are integer multiples of thefundamental frequency of the waveform of the harmonically rich signal608. These sinusoidal waves are referred to as the harmonics of theunderlying waveform, and the fundamental frequency is referred to as thefirst harmonic. FIG. 6F and FIG. 6G show separately the sinusoidalcomponents making up the first, third, and fifth harmonics of section610 and section 612. (Note that in theory there may be an infinitenumber of harmonics; in this example, because harmonically rich signal608 is shown as a square wave, there are only odd harmonics). Threeharmonics are shown simultaneously (but not summed) in FIG. 6H.

[0309] The relative amplitudes of the harmonics are generally a functionof the relative widths of the pulses of harmonically rich signal 306 andthe period of the fundamental frequency, and can be determined by doinga Fourier analysis of harmonically rich signal 306. According to anembodiment of the invention, the input signal 606 may be shaped toensure that the amplitude of the desired harmonic is sufficient for itsintended use (e.g., transmission).

[0310] A filter 308 filters out any undesired frequencies (harmonics),and outputs an electromagnetic (EM) signal at the desired harmonicfrequency or frequencies as an output signal 310, shown for example as afiltered output signal 614 in FIG. 61.

[0311]FIG. 4 illustrates an example universal frequency up-conversion(UFU) module 401. The UFU module 401 includes an example switch module304, which comprises a bias signal 402, a resistor or impedance 404, auniversal frequency translator (UFT) 450, and a ground 408. The UFT 450includes a switch 406. The input signal 302 (designated as “ControlSignal” in FIG. 4) controls the switch 406 in the UFT 450, and causes itto close and open. Harmonically rich signal 306 is generated at a node405 located between the resistor or impedance 404 and the switch 406.

[0312] Also in FIG. 4, it can be seen that an example filter 308 iscomprised of a capacitor 410 and an inductor 412 shunted to a ground414. The filter is designed to filter out the undesired harmonics ofharmonically rich signal 306.

[0313] The invention is not limited to the UFU embodiment shown in FIG.4.

[0314] For example, in an alternate embodiment shown in FIG. 5, anunshaped input signal 501 is routed to a pulse shaping module 502. Thepulse shaping module 502 modifies the unshaped input signal 501 togenerate a (modified) input signal 302 (designated as the “ControlSignal” in FIG. 5). The input signal 302 is routed to the switch module304, which operates in the manner described above. Also, the filter 308of FIG. 5 operates in the manner described above.

[0315] The purpose of the pulse shaping module 502 is to define thepulse width of the input signal 302. Recall that the input signal 302controls the opening and closing of the switch 406 in switch module 304.During such operation, the pulse width of the input signal 302establishes the pulse width of the harmonically rich signal 306. Asstated above, the relative amplitudes of the harmonics of theharmonically rich signal 306 are a function of at least the pulse widthof the harmonically rich signal 306. As such, the pulse width of theinput signal 302 contributes to setting the relative amplitudes of theharmonics of harmonically rich signal 306.

[0316] Further details of up-conversion as described in this section arepresented in pending U.S. application “Method and System for FrequencyUp-Conversion,” Ser. No. 09/176,154, filed Oct. 21, 1998, incorporatedherein by reference in its entirety.

[0317] 5. Enhanced Signal Reception

[0318] The present invention is directed to systems and methods ofenhanced signal reception (ESR), and applications of same. Referring toFIG. 21, transmitter 2104 accepts a modulating baseband signal 2102 andgenerates (transmitted) redundant spectrums 2106 a-n, which are sentover communications medium 2108. Receiver 2112 recovers a demodulatedbaseband signal 2114 from (received) redundant spectrums 2110 a-n.Demodulated baseband signal 2114 is representative of the modulatingbaseband signal 2102, where the level of similarity between themodulating baseband signal 2114 and the modulating baseband signal 2102is application dependent.

[0319] Modulating baseband signal 2102 is preferably any informationsignal desired for transmission and/or reception. An example modulatingbaseband signal 2202 is illustrated in FIG. 22A, and has an associatedmodulating baseband spectrum 2204 and image spectrum 2203 that areillustrated in FIG. 22B. Modulating baseband signal 2202 is illustratedas an analog signal in FIG. 22a, but could also be a digital signal, orcombination thereof. Modulating baseband signal 2202 could be a voltage(or current) characterization of any number of real world occurrences,including for example and without limitation, the voltage (or current)representation for a voice signal.

[0320] Each transmitted redundant spectrum 2106 a-n contains thenecessary information to substantially reconstruct the modulatingbaseband signal 2102. In other words, each redundant spectrum 2106 a-ncontains the necessary amplitude, phase, and frequency information toreconstruct the modulating baseband signal 2102.

[0321]FIG. 22C illustrates example transmitted redundant spectrums 2206b-d. Transmitted redundant spectrums 2206 b-d are illustrated to containthree redundant spectrums for illustration purposes only. Any number ofredundant spectrums could be generated and transmitted as will beexplained in following discussions.

[0322] Transmitted redundant spectrums 2206 b-d are centered at f₁, witha frequency spacing f₂ between adjacent spectrums. Frequencies f₁ and f₂are dynamically adjustable in real-time as will be shown below. FIG. 22Dillustrates an alternate embodiment, where redundant spectrums 2208 c,dare centered on unmodulated oscillating signal 2209 at f₁ (Hz).Oscillating signal 2209 may be suppressed if desired using, for example,phasing techniques or filtering techniques. Transmitted redundantspectrums are preferably above baseband frequencies as is represented bybreak 2205 in the frequency axis of FIGS. 22C and 22D.

[0323] Received redundant spectrums 2110 a-n are substantially similarto transmitted redundant spectrums 2106 a-n, except for the changesintroduced by the communications medium 2108. Such changes can includebut are not limited to signal attenuation, and signal interference. FIG.22E illustrates example received redundant spectrums 2210 b-d. Receivedredundant spectrums 2210 b-d are substantially similar to transmittedredundant spectrums 2206 b-d, except that redundant spectrum 2210 cincludes an undesired jamming signal spectrum 2211 in order toillustrate some advantages of the present invention. Jamming signalspectrum 2211 is a frequency spectrum associated with a jamming signal.For purposes of this invention, a “jamming signal” refers to anyunwanted signal, regardless of origin, that may interfere with theproper reception and reconstruction of an intended signal. Furthermore,the jamming signal is not limited to tones as depicted by spectrum 2211,and can have any spectral shape, as will be understood by those skilledin the art(s).

[0324] As stated above, demodulated baseband signal 2114 is extractedfrom one or more of received redundant spectrums 2210 b-d. FIG. 22Fillustrates example demodulated baseband signal 2212 that is, in thisexample, substantially similar to modulating baseband signal 2202 (FIG.22A); where in practice, the degree of similarity is applicationdependent.

[0325] An advantage of the present invention should now be apparent. Therecovery of modulating baseband signal 2202 can be accomplished byreceiver 2112 in spite of the fact that high strength jamming signal(s)(e.g. jamming signal spectrum 2211) exist on the communications medium.The intended baseband signal can be recovered because multiple redundantspectrums are transmitted, where each redundant spectrum carries thenecessary information to reconstruct the baseband signal. At thedestination, the redundant spectrums are isolated from each other sothat the baseband signal can be recovered even if one or more of theredundant spectrums are corrupted by a jamming signal.

[0326] Transmitter 2104 will now be explored in greater detail. FIG. 23Aillustrates transmitter 2301, which is one embodiment of transmitter2104 that generates redundant spectrums configured similar to redundantspectrums 2206 b-d. Transmitter 2301 includes generator 2303, optionalspectrum processing module 2304, and optional medium interface module2320. Generator 2303 includes: first oscillator 2302, secondoscillator2309, first stage modulator 2306, and second stage modulator2310.

[0327] Transmitter 2301 operates as follows. First oscillator 2302 andsecond oscillator 2309 generate a first oscillating signal 2305 andsecond oscillating signal 2312, respectively. First stage modulator 2306modulates first oscillating signal 2305 with modulating baseband signal2202, resulting in modulated signal 2308. First stage modulator 2306 mayimplement any type of modulation including but not limited to: amplitudemodulation, frequency modulation, phase modulation, combinationsthereof, or any other type of modulation. Second stage modulator 2310modulates modulated signal 2308 with second oscillating signal 2312,resulting in multiple redundant spectrums 2206 a-n shown in FIG. 23B.Second stage modulator 2310 is preferably a phase modulator, or afrequency modulator, although other types of modulation may beimplemented including but not limited to amplitude modulation. Eachredundant spectrum 2206 a-n contains the necessary amplitude, phase, andfrequency information to substantially reconstruct the modulatingbaseband signal 2202.

[0328] Redundant spectrums 2206 a-n are substantially centered aroundf₁, which is the characteristic frequency of first oscillating signal2305. Also, each redundant spectrum 2206 a-n (except for 2206 c) isoffset from f, by approximately a multiple of f₂ (Hz), where f₂ is thefrequency of the second oscillating signal 2312. Thus, each redundantspectrum 2206 a-n is offset from an adjacent redundant spectrum by f₂(Hz). This allows the spacing between adjacent redundant spectrums to beadjusted (or tuned) by changing f₂ that is associated with secondoscillator 2309. Adjusting the spacing between adjacent redundantspectrums allows for dynamic real-time tuning of the bandwidth occupiedby redundant spectrums 2206 a-n.

[0329] In one embodiment, the number of redundant spectrums 2206 a-ngenerated by transmitter 2301 is arbitrary and may. be unlimited asindicated by the “a-n” designation for redundant spectrums 2206 a-n.However, a typical communications medium will have a physical and/oradministrative limitations (i.e. FCC regulations) that restrict thenumber of redundant spectrums that can be practically transmitted overthe communications medium. Also, there may be other reasons to limit thenumber of redundant spectrums transmitted. Therefore, preferably, thetransmitter 2301 will include an optional spectrum processing module2304 to process the redundant spectrums 2206 a-n prior to transmissionover communications medium 2108.

[0330] In one embodiment, spectrum processing module 2304 includes afilter with a passband 2207 (FIG. 23C) to select redundant spectrums2206 b-d for transmission. This will substantially limit the frequencybandwidth occupied by the redundant spectrums to the passband 2207. Inone embodiment, spectrum processing module 2304 also up convertsredundant spectrums and/or amplifies redundant spectrums prior totransmission over the communications medium 2108. Finally, mediuminterface module 2320 transmits redundant spectrums over thecommunications medium 2108. In one embodiment, communications medium2108 is an over-the-air link and medium interface module 2320 is anantenna. Other embodiments for communications medium 2108 and mediuminterface module 2320 will be understood based on the teachingscontained herein.

[0331]FIG. 23D illustrates transmitter 2321, which is one embodiment oftransmitter 2104 that generates redundant spectrums configured similarto redundant spectrums 2208 c-d and unmodulated spectrum 2209.Transmitter 2321 includes generator 2311, spectrum processing module2304, and (optional) medium interface module 2320. Generator 2311includes: first oscillator 2302, second oscillator 2309, first stagemodulator 2306, and second stage modulator 2310.

[0332] As shown in FIG. 23D, many of the components in transmitter 2321are similar to those in transmitter 2301. However, in this embodiment,modulating baseband signal 2202 modulates second oscillating signal2312. Transmitter 2321 operates as follows. First stage modulator 2306modulates second oscillating signal 2312 with modulating baseband signal2202, resulting in modulated signal 2322. As described earlier, firststage modulator 2306 can effect any type of modulation including but notlimited to: amplitude modulation frequency modulation, combinationsthereof, or any other type of modulation. Second stage modulator 2310modulates first oscillating signal 2304 with modulated signal 2322,resulting in redundant spectrums 2208 a-n, as shown in FIG. 23E. Secondstage modulator 2310 is preferably a phase or frequency modulator,although other modulators could used including but not limited to anamplitude modulator.

[0333] Redundant spectrums 2208 a-n are centered on unmodulated spectrum2209 (at f₁ Hz), and adjacent spectrums are separated by f₂ Hz. Thenumber of redundant spectrums 2208 a-n generated by generator 2311 isarbitrary and unlimited, similar to spectrums 2206 a-n discussed above.Therefore, optional spectrum processing module 2304 may also include afilter with passband 2325 to select, for example, spectrums 2208c,d fortransmission over communications medium 2108. In addition, optionalspectrum processing module 2304 may also include a filter (such as abands top filter) to attenuate unmodulated spectrum 2209. Alternatively,unmodulated spectrum 2209 may be attenuated by using phasing techniquesduring redundant spectrum generation. Finally, (optional) mediuminterface module 2320 transmits redundant spectrums 2208 c,d overcommunications medium 2108.

[0334] Receiver 2112 will now be explored in greater detail toillustrate recovery of a demodulated baseband signal from receivedredundant spectrums. FIG. 24A illustrates receiver 2430, which is oneembodiment of receiver 2112. Receiver 2430 includes optional mediuminterface module 2402, down-converter 2404, spectrum isolation module2408, and data extraction module 2414. Spectrum isolation module 2408includes filters 2410 a-c. Data extraction module 2414 includesdemodulators 2416 a-c, error check modules 2420 a-c, and arbitrationmodule 2424. Receiver 2430 will be discussed in relation to the signaldiagrams in FIGS. 24B-24J.

[0335] In one embodiment, optional medium interface module 2402 receivesredundant spectrums 2210 b-d (FIG. 22E, and FIG. 24B). Each redundantspectrum 2210 b-d includes the necessary amplitude, phase, and frequencyinformation to substantially reconstruct the modulating baseband signalused to generated the redundant spectrums. However, in the presentexample, spectrum 2210 c also contains jamming signal 2211, which mayinterfere with the recovery of a baseband signal from spectrum 2210 c.Down-converter 2404 down-converts received redundant spectrums 2210 b-dto lower intermediate frequencies, resulting in redundant spectrums 2406a-c (FIG. 24C). Jamming signal 2211 is also down-converted to jammingsignal 2407, as it is contained within redundant spectrum 2406 b.Spectrum isolation module 2408 includes filters 2410a-c that isolateredundant spectrums 2406 a-c from each other (FIGS. 24D-24F,respectively). Demodulators 2416 a-c independently demodulate spectrums2406 a-c, resulting in demodulated baseband signals 2418 a-c,respectively (FIGS. 24G-241). Error check modules 2420 a-c analyzedemodulate baseband signal 2418 a-c to detect any errors. In oneembodiment, each error check module 2420a-c sets an error flag 2422 a-cwhenever an error is detected in a demodulated baseband signal.Arbitration module 2424 accepts the demodulated baseband signals andassociated error flags, and selects a substantially error-freedemodulated baseband signal (FIG. 24J). In one embodiment, thesubstantially error-free demodulated baseband signal will besubstantially similar to the modulating baseband signal used to generatethe received redundant spectrums, where the degree of similarity isapplication dependent.

[0336] Referring to FIGS. 24G-I, arbitration module 2424 will selecteither demodulated baseband signal 2418 a or 2418 c, because error checkmodule 2420 b will set the error flag 2422 b that is associated withdemodulated baseband signal 2418 b.

[0337] The error detection schemes implemented by the error detectionmodules include but are not limited to: cyclic redundancy check (CAC)and parity check for digital signals, and various error detectionsschemes for analog signal.

[0338] Further details of enhanced signal reception as described in thissection are presented in pending U.S. application “Method and System forEnsuring Reception of a Communications Signal,” Ser. No. 09/176,415,filed Oct. 21, 1998, incorporated herein by reference in its entirety.

[0339] 6. Unified Down-Conversion and Filtering

[0340] The present invention is directed to systems and methods ofunified down-conversion and filtering (UDF), and applications of same.

[0341] In particular, the present invention includes a unifieddown-converting and filtering (UDF) module that performs frequencyselectivity and frequency translation in a unified (i.e., integrated)manner. By operating in this manner, the invention achieves highfrequency selectivity prior to frequency translation (the invention isnot limited to this embodiment). The invention achieves high frequencyselectivity at substantially any frequency, including but not limited toRF (radio frequency) and greater frequencies. It should be understoodthat the invention is not limited to this example of RF and greaterfrequencies. The invention is intended, adapted, and capable of workingwith lower than radio frequencies.

[0342]FIG. 17 is a conceptual block diagram of a UDF module 1702according to an embodiment of the present invention. The UDF module 1702performs at least frequency translation and frequency selectivity.

[0343] The effect achieved by the UDF module 1702 is to perform thefrequency selectivity operation prior to the performance of thefrequency translation operation. Thus, the UDF module 1702 effectivelyperforms input filtering.

[0344] According to embodiments of the present invention, such inputfiltering involves a relatively narrow bandwidth. For example, suchinput filtering may represent channel select filtering, where the filterbandwidth may be, for example, 50 KHz to 150 KHz. It should beunderstood, however, that the invention is not limited to thesefrequencies. The invention is intended, adapted, and capable ofachieving filter bandwidths of less than and greater than these values.

[0345] In embodiments of the invention, input signals 1704 received bythe UDF module 1702 are at radio frequencies. The UDF module 1702effectively operates to input filter these RF input signals 1704.Specifically, in these embodiments, the UDF module 1702 effectivelyperforms input, channel select filtering of the RF input signal 1704.Accordingly, the invention achieves high selectivity at highfrequencies.

[0346] The UDF module 1702 effectively performs various types offiltering, including but not limited to bandpass filtering, low passfiltering, high pass filtering, notch filtering, all pass filtering,band stop filtering, etc., and combinations thereof.

[0347] Conceptually, the UDF module 1702 includes a frequency translator1708. The frequency translator 1708 conceptually represents that portionof the UDF module 1702 that performs frequency translation (downconversion).

[0348] The UDF module 1702 also conceptually includes an apparent inputfilter 1706 (also sometimes called an input filtering emulator).Conceptually, the apparent input filter 1706 represents that portion ofthe UDF module 1702 that performs input filtering.

[0349] In practice, the input filtering operation performed by the UDFmodule 1702 is integrated with the frequency translation operation. Theinput filtering operation can be viewed as being performed concurrentlywith the frequency translation operation. This is a reason why the inputfilter 1706 is herein referred to as an “apparent” input filter 1706.

[0350] The UDF module 1702 of the present invention includes a number ofadvantages. For example, high selectivity at high frequencies isrealizable using the UDF module 1702. This feature of the invention isevident by the high Q factors that are attainable. For example, andwithout limitation, the UDF module 1702 can be designed with a filtercenter frequency f_(C) on the order of 900 MHZ, and a filter bandwidthon the order of 50 KHz. This represents a Q of 18,000 (Q is equal to thecenter frequency divided by the bandwidth).

[0351] It should be understood that the invention is not limited tofilters with high Q factors. The filters contemplated by the presentinvention may have lesser or greater Qs, depending on the application,design, and/or implementation. Also, the scope of the invention includesfilters where Q factor as discussed herein is not applicable.

[0352] The invention exhibits additional advantages. For example, thefiltering center frequency f_(C) of the UDF module 1702 can beelectrically adjusted, either statically or dynamically.

[0353] Also, the UDF module 1702 can be designed to amplify inputsignals.

[0354] Further, the UDF module 1702 can be implemented without largeresistors, capacitors, or inductors. Also, the UDF module 1702 does notrequire that tight tolerances be maintained on the values of itsindividual components, i.e., its resistors, capacitors, inductors, etc.As a result, the architecture of the UDF module 1702 is friendly tointegrated circuit design techniques and processes.

[0355] The features and advantages exhibited by the UDF module 1702 areachieved at least in part by adopting a new technological paradigm withrespect to frequency selectivity and translation. Specifically,according to the present invention, the UDF module 1702 performs thefrequency selectivity operation and the frequency translation operationas a single, unified (integrated) operation. According to the invention,operations relating to frequency translation also contribute to theperformance of frequency selectivity, and vice versa.

[0356] According to embodiments of the present invention, the UDF modulegenerates an output signal from an input signal using samples/instancesof the input signal and samples/instances of the output signal.

[0357] More particularly, first, the input signal is under-sampled. Thisinput sample includes information (such as amplitude, phase, etc.)representative of the input signal existing at the time the sample wastaken.

[0358] As described further below, the effect of repetitively performingthis step is to translate the frequency (that is, down-convert) of theinput signal to a desired lower frequency, such as an intermediatefrequency (IF) or baseband.

[0359] Next, the input sample is held (that is, delayed).

[0360] Then, one or more delayed input samples (some of which may havebeen scaled) are combined with one or more delayed instances of theoutput signal (some of which may have been scaled) to generate a currentinstance of the output signal.

[0361] Thus, according to a preferred embodiment of the invention, theoutput signal is generated from prior samples/instances of the inputsignal and/or the output signal. (It is noted that, in some embodimentsof the invention, current samples/instances of the input signal and/orthe output signal may be used to generate current instances of theoutput signal.). By operating in this manner, the UDF module preferablyperforms input filtering and frequency down-conversion in a unifiedmanner.

[0362]FIG. 19 illustrates an example implementation of the unifieddown-converting and filtering (UDF) module 1922. The UDF module 1922performs the frequency translation operation and the frequencyselectivity operation in an integrated, unified manner as describedabove, and as further described below.

[0363] In the example of FIG. 19, the frequency selectivity operationperformed by the UDF module 1922 comprises a band-pass filteringoperation according to EQ. 1, below, which is an example representationof a band-pass filtering transfer function.

VO=α ₁ z ³¹ ¹ VI−β ₁ z ⁻¹ VO−β ₀ z ⁻² VO  EQ. 1

[0364] It should be noted, however, that the invention is not limited toband-pass filtering. Instead, the invention effectively performs varioustypes of filtering, including but not limited to bandpass filtering, lowpass filtering, high pass filtering, notch filtering, all passfiltering, band stop filtering, etc., and combinations thereof. As willbe appreciated, there are many representations of any given filter type.The invention is applicable to these filter representations. Thus, EQ. 1is referred to herein for illustrative purposes only, and is notlimiting.

[0365] The UDF module 1922 includes a down-convert and delay module1924, first and second delay modules 1928 and 1930, first and secondscaling modules 1932 and 1934, an output sample and hold module 1936,and an (optional) output smoothing module 1938. Other embodiments of theUDF module will have these components in different configurations,and/or a subset of these components, and/or additional components. Forexample, and without limitation, in the configuration shown in FIG. 19,the output smoothing module 1938 is optional.

[0366] As further described below, in the example of FIG. 19, thedown-convert and delay module 1924 and the first and second delaymodules 1928 and 1930 include switches that are controlled by a clockhaving two phases, φ₁ and φ₂. φ₁ and φ₂ preferably have the samefrequency, and are non-overlapping (alternatively, a plurality such astwo clock signals having these characteristics could be used). As usedherein, the term “non-overlapping” is defined as two or more signalswhere only one of the signals is active at any given time. In someembodiments, signals are “active” when they are high. In otherembodiments, signals are active when they are low.

[0367] Preferably, each of these switches closes on a rising edge of φ₁or φ₂, and opens on the next corresponding falling edge of φ₁ or φ₂.However, the invention is not limited to this example. As will beapparent to persons skilled in the relevant art(s), other clockconventions can be used to control the switches.

[0368] In the example of FIG. 19, it is assumed that α₁ is equal to one.Thus, the output of the down-convert and delay module 1924 is notscaled. As evident from the embodiments described above, however, theinvention is not limited to this example.

[0369] The example UDF module 1922 has a filter center frequency of900.2 MHZ and a filter bandwidth of 570 KHz. The pass band of the UDFmodule 1922 is on the order of 899.915 MHZ to 900.485 MHZ. The Q factorof the UDF module 1922 is approximately 1879 (i.e., 900.2 MHZ divided by570 KHz).

[0370] The operation of the UDF module 1922 shall now be described withreference to a Table 1802 (FIG. 18) that indicates example values atnodes in the UDF module 1922 at a number of consecutive time increments.It is assumed in Table 1802 that the UDF module 1922 begins operating attime t−1. As indicated below, the UDF module 1922 reaches steady state afew time units after operation begins. The number of time unitsnecessary for a given UDF module to reach steady state depends on theconfiguration of the UDF module, and will be apparent to persons skilledin the relevant art(s) based on the teachings contained herein.

[0371] At the rising edge of φ₁ at time t−1, a switch 1950 in thedown-convert and delay module 1924 closes. This allows a capacitor 1952to charge to the current value of an input signal, VI_(t−1), such thatnode 1902 is at VI_(t−1). This is indicated by cell 1804 in FIG. 18. Ineffect, the combination of the switch 1950 and the capacitor 1952 in thedown-convert and delay module 1924 operates to translate the frequencyof the input signal VI to a desired lower frequency, such as IF orbaseband. Thus, the value stored in the capacitor 1952 represents aninstance of a down-converted image of the input signal VI.

[0372] The manner in which the down-convert and delay module 1924performs frequency down-conversion is further described elsewhere inthis application, and is additionally described in pending U.S.application “Method and System for Down-Converting ElectromagneticSignals,” Ser. No. 09/176,022, filed Oct. 21, 1998, which is hereinincorporated by reference in its entirety.

[0373] Also at the rising edge of φ₁ at time t−1, a switch 1958 in thefirst delay module 1928 closes, allowing a capacitor 1960 to charge toVO_(t−1), such that node 1906 is at VO_(t−1). This is indicated by cell1806 in Table 1802. (In practice, VO_(t−1) is undefined at this point.However, for ease of understanding, VO_(t−1) shall continue to be usedfor purposes of explanation.)

[0374] Also at the rising edge of φ₁ at time t−1, a switch 1966 in thesecond delay module 1930 closes, allowing a capacitor 1968 to charge toa value stored in a capacitor 1964. At this time, however, the value incapacitor 1964 is undefined, so the value in capacitor 1968 isundefined. This is indicated by cell 1807 in table 1802.

[0375] At the rising edge of φ₂ at time t−1, a switch 1954 in thedown-convert and delay module 1924 closes, allowing a capacitor 1956 tocharge to the level of the capacitor 1952. Accordingly, the capacitor1956 charges to VI_(t−1), such that node 1904 is at VI_(t−1). This isindicated by cell 1810 in Table 1802.

[0376] The UDF module 1922 may optionally include a unity gain module1990A between capacitors 1952 and 1956. The unity gain module 1990Aoperates as a current source to enable capacitor 1956 to charge withoutdraining the charge from capacitor 1952. For a similar reason, the UDFmodule 1922 may include other unity gain modules 1990B-1990G. It shouldbe understood that, for many embodiments and applications of theinvention, these unity gain modules 1990A-1990G are optional. Thestructure and operation of the unity gain modules 1990 will be apparentto persons skilled in the relevant art(s).

[0377] Also at the rising edge of φ₂ at time t−1, a switch 1962 in thefirst delay module 1928 closes, allowing a capacitor 1964 to charge tothe level of the capacitor 1960. Accordingly, the capacitor 1964 chargesto VO_(t−1), such that node 1908 is at VO_(t−1). This is indicated bycell 1814 in Table 1802.

[0378] Also at the rising edge of φ₂ at time t−1, a switch 1970 in thesecond delay module 1930 closes, allowing a capacitor 1972 to charge toa value stored in a capacitor 1968. At this time, however, the value incapacitor 1968 is undefined, so the value in capacitor 1972 isundefined. This is indicated by cell 1815 in table 1802.

[0379] At time t, at the rising edge of φ₁, the switch 1950 in thedown-convert and delay module 1924 closes. This allows the capacitor1952 to charge to VI_(t−1), such that node 1902 is at VI_(t). This isindicated in cell 1816 of Table 1802.

[0380] Also at the rising edge of φ₁ at time t, the switch 1958 in thefirst delay module 1928 closes, thereby allowing the capacitor 1960 tocharge to VO_(t). Accordingly, node 1906 is at VO_(t). This is indicatedin cell 1820 in Table 1802.

[0381] Further at the rising edge of φ₁ at time t, the switch 1966 inthe second delay module 1930 closes, allowing a capacitor 1968 to chargeto the level of the capacitor 1964. Therefore, the capacitor 1968charges to VO_(t−1), such that node 1910 is at VO_(t−1). This isindicated by cell 1824 in Table 1802.

[0382] At the rising edge of φ₂ at time t, the switch 1954 in thedown-convert and delay module 1924 closes, allowing the capacitor 1956to charge to the level of the capacitor 1952. Accordingly, the capacitor1956 charges to VI_(t), such that node 1904 is at VI_(t). This isindicated by cell 1828 in Table 1802.

[0383] Also at the rising edge of φ₂ at time t, the switch 1962 in thefirst delay module 1928 closes, allowing the capacitor 1964 to charge tothe level in the capacitor 1960. Therefore, the capacitor 1964 chargesto VO_(t), such that node 1908 is at VO_(t). This is indicated by cell1832 in Table 1802.

[0384] Further at the rising edge of φ₂ at time t, the switch 1970 inthe second delay module 1930 closes, allowing the capacitor 1972 in thesecond delay module 1930 to charge to the level of the capacitor 1968 inthe second delay module 1930. Therefore, the capacitor 1972 charges toVO_(t−1), such that node 1912 is at VO_(t−1). This is indicated in cell1836 of FIG. 18.

[0385] At time t+1, at the rising edge of φ₁, the switch 1950 in thedown-convert and delay module 1924 closes, allowing the capacitor 1952to charge to VI_(t+1). Therefore, node 1902 is at VI_(t+1), as indicatedby cell 1838 of Table 1802.

[0386] Also at the rising edge of φ₁ at time t+1, the switch 1958 in thefirst delay module 1928 closes, allowing the capacitor 1960 to charge toVO_(t+1). Accordingly, node 1906 is at VO_(t+1), as indicated by cell1842 in Table 1802.

[0387] Further at the rising edge of φ₁, at time t+1, the switch 1966 inthe second delay module 1930 closes, allowing the capacitor 1968 tocharge to the level of the capacitor 1964. Accordingly, the capacitor1968 charges to VO_(t), as indicated by cell 1846 of Table 1802.

[0388] In the example of FIG. 19, the first scaling module 1932 scalesthe value at node 1908 (i.e., the output of the first delay module 1928)by a scaling factor of −0.1. Accordingly, the value present at node 1914at time t+1 is −0.1* VO_(t). Similarly, the second scaling module 1934scales the value present at node 1912 (i.e., the output of the secondscaling module 1930) by a scaling factor of −0.8. Accordingly, the valuepresent at node 1916 is −0.8*VO_(t−1), at time t+1.

[0389] At time t+1, the values at the inputs of the summer 1926 are:VI_(t) at node 1904, −0.1*VO_(t) at node 1914, and −0.8* VO_(t−1) atnode 1916 (in the example of FIG. 19, the values at nodes 1914 and 1916are summed by a second summer 1925, and this sum is presented to thesummer 1926). Accordingly, at time t+1, the summer generates a signalequal to VI_(t)−0.1*VO_(t)−0.8* VO_(t−1).

[0390] At the rising edge of φ₁ at time t+1, a switch 1991 in the outputsample and hold module 1936 closes, thereby allowing a capacitor 1992 tocharge to VO_(t+1). Accordingly, the capacitor 1992 charges to VO_(t+1),which is equal to the sum generated by the adder 1926. As just noted,this value is equal to: VI_(t)−0.1*VO_(t)−0.8*VO_(t−1). This isindicated in cell 1850 of Table 1802. This value is presented to theoptional output smoothing module 1938, which smooths the signal tothereby generate the instance of the output signal VO_(t+1). It isapparent from inspection that this value of VO_(t+1) is consistent withthe band pass filter transfer function of EQ. 1.

[0391] Further details of unified down-conversion and filtering asdescribed in this section are presented in pending U.S. application“Integrated Frequency Translation And Selectivity,” Ser. No. 09/175,966,filed Oct. 21, 1998, incorporated herein by reference in its entirety.

[0392] 7. Example Application Embodiments of the Invention

[0393] As noted above, the UFT module of the present invention is a verypowerful and flexible device. Its flexibility is illustrated, in part,by the wide range of applications in which it can be used. Its power isillustrated, in part, by the usefulness and performance of suchapplications.

[0394] Example applications of the UFT module were described above. Inparticular, frequency down-conversion, frequency up-conversion, enhancedsignal reception, and unified down-conversion and filtering applicationsof the UFT module were summarized above, and are further describedbelow. These applications of the UFT module are discussed herein forillustrative purposes. The invention is not limited to these exampleapplications. Additional applications of the UFT module will be apparentto persons skilled in the relevant art(s), based on the teachingscontained herein.

[0395] For example, the present invention can be used in applicationsthat involve frequency down-conversion. This is shown in FIG. 1C, forexample, where an example UFT module 115 is used in a down-conversionmodule 114. In this capacity, the UFT module 115 frequency down-convertsan input signal to an output signal. This is also shown in FIG. 7, forexample, where an example UFT module 706 is part of a down-conversionmodule 704, which is part of a receiver 702.

[0396] The present invention can be used in applications that involvefrequency up-conversion. This is shown in FIG. 1D, for example, where anexample UFT module 117 is used in a frequency up-conversion module 116.In this capacity, the UFT module 117 frequency up-converts an inputsignal to an output signal. This is also shown in FIG. 8, for example,where an example UFT module 806 is part of up-conversion module 804,which is part of a transmitter 802.

[0397] The present invention can be used in environments having one ormore transmitters 902 and one or more receivers 906, as illustrated inFIG. 9. In such environments, one or more of the transmitters 902 may beimplemented using a UFT module, as shown for example in FIG. 8. Also,one or more of the receivers 906 may be implemented using a UFT module,as shown for example in FIG. 7.

[0398] The invention can be used to implement a transceiver. An exampletransceiver 1002 is illustrated in FIG. 10. The transceiver 1002includes a transmitter 1004 and a receiver 1008. Either the transmitter1004 or the receiver 1008 can be implemented using a UFT module.Alternatively, the transmitter 1004 can be implemented using a UFTmodule 1006, and the receiver 1008 can be implemented using a UFT module1010. This embodiment is shown in FIG. 10.

[0399] Another transceiver embodiment according to the invention isshown in FIG. 11. In this transceiver 1102, the transmitter 1104 and thereceiver 1108 are implemented using a single UFT module 1106. In otherwords, the transmitter 1104 and the receiver 1108 share a UFT module1106.

[0400] As described elsewhere in this application, the invention isdirected to methods and systems for enhanced signal reception (ESR).Various ESR embodiments include an ESR module (transmit) in atransmitter 1202, and an ESR module (receive) in a receiver 1210. Anexample ESR embodiment configured in this manner is illustrated in FIG.12.

[0401] The ESR module (transmit) 1204 includes a frequency up-conversionmodule 1206. Some embodiments of this frequency up-conversion module1206 may be implemented using a UFT module, such as that shown in FIG.1D.

[0402] The ESR module (receive) 1212 includes a frequencydown-conversion module 1214. Some embodiments of this frequencydown-conversion module 1214 may be implemented using a UFT module, suchas that shown in FIG. 1C.

[0403] As described elsewhere in this application, the invention isdirected to methods and systems for unified down-conversion andfiltering (UDF). An example unified down-conversion and filtering module1302 is illustrated in FIG. 13. The unified down-conversion andfiltering module 1302 includes a frequency down-conversion module 1304and a filtering module 1306. According to the invention, the frequencydown-conversion module 1304 and the filtering module 1306 areimplemented using a UFT module 1308, as indicated in FIG. 13.

[0404] Unified down-conversion and filtering according to the inventionis useful in applications involving filtering and/or frequencydown-conversion. This is depicted, for example, in FIGS. 15A-15F. FIGS.15A-15C indicate that unified down-conversion and filtering according tothe invention is useful in applications where filtering precedes,follows, or both precedes and follows frequency down-conversion. FIG.15D indicates that a unified down-conversion and filtering module 1524according to the invention can be utilized as a filter 1522 (i.e., wherethe extent of frequency down-conversion by the down-converter in theunified down-conversion and filtering module 1524 is minimized). FIG.15E indicates that a unified down-conversion and filtering module 1528according to the invention can be utilized as a down-converter 1526(i.e., where the filter in the unified down-conversion and filteringmodule 1528 passes substantially all frequencies). FIG. 15F illustratesthat the unified down-conversion and filtering module 1532 can be usedas an amplifier. It is noted that one or more UDF modules can be used inapplications that involve at least one or more of filtering, frequencytranslation, and amplification.

[0405] For example, receivers, which typically perform filtering,down-conversion, and filtering operations, can be implemented using oneor more unified down-conversion and filtering modules. This isillustrated, for example, in FIG. 14.

[0406] The methods and systems of unified down-conversion and filteringof the invention have many other applications. For example, as discussedherein, the enhanced signal reception (ESR) module (receive) operates todown-convert a signal containing a plurality of spectrums. The ESRmodule (receive) also operates to isolate the spectrums in thedown-converted signal, where such isolation is implemented via filteringin some embodiments. According to embodiments of the invention, the ESRmodule (receive) is implemented using one or more unifieddown-conversion and filtering (UDF) modules. This is illustrated, forexample, in FIG. 16. In the example of FIG. 16, one or more of the UDFmodules 1610, 1612, 1614 operates to down-convert a received signal. TheUDF modules 1610, 1612, 1614 also operate to filter the down-convertedsignal so as to isolate the spectrum(s) contained therein. As notedabove, the UDF modules 1610, 1612, 1614 are implemented using theuniversal frequency translation (UFT) modules of the invention.

[0407] The invention is not limited to the applications of the UFTmodule described above. For example, and without limitation, subsets ofthe applications (methods and/or structures) described herein (andothers that would be apparent to persons skilled in the relevant art(s)based on the herein teachings) can be associated to form usefulcombinations.

[0408] For example, transmitters and receivers are two applications ofthe UFT module. FIG. 10 illustrates a transceiver 1002 that is formed bycombining these two applications of the UFT module, i.e., by combining atransmitter 1004 with a receiver 1008.

[0409] Also, ESR (enhanced signal reception) and unified down-conversionand filtering are two other applications of the UFT module. FIG. 16illustrates an example where ESR and unified down-conversion andfiltering are combined to form a modified enhanced signal receptionsystem.

[0410] The invention is not limited to the example applications of theUFT module discussed herein. Also, the invention is not limited to theexample combinations of applications of the UFT module discussed herein.These examples were provided for illustrative purposes only, and are notlimiting. Other applications and combinations of such applications willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. Such applications and combinations include,for example and without limitation, applications/combinations comprisingand/or involving one or more of: (1) frequency translation; (2)frequency down-conversion; (3) frequency up-conversion; (4) receiving;(5) transmitting; (6) filtering; and/or (7) signal transmission andreception in environments containing potentially jamming signals.

[0411] Additional examples are set forth below describing applicationsof the UFT module with circuits that reduce or eliminate unwanted DCoffset and re-radiation, and improve dynamic range.

[0412] 7.0 DC Offset, Re-Radiation, and Dynamic Range Considerations andCorrections

[0413] Various embodiments related to the method(s) and structure(s)described herein are presented in this section (and its subsections).Problems related to DC offset, re-radiation, and dynamic range aredescribed below. Applications of the UFT module are provided in relationto circuits used to reduce or eliminate problems of DC offset andre-radiation, and to improve dynamic range.

[0414] These embodiments are described herein for purposes ofillustration, and not limitation. The invention is not limited to theseembodiments. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. The invention is intended and adapted toinclude such alternate embodiments.

[0415] 7.1 Overview of DC Offset and Re-Radiation

[0416] Receivers, and other electronic circuits, may suffer fromproblems of DC offset and re-radiation. Generally, “DC offset” refers toa DC voltage level that is added to a signal of interest by relatedcircuitry. The related circuitry creates the DC offset voltage through avariety of mechanisms that are well known. Some of these mechanisms arediscussed in further detail below. If a DC offset voltage value issignificant, it can degrade the quality of the signal of interest. In areceiver, for example, the signal of interest may be a down-convertedsignal. Unless reduced or eliminated, the added DC offset voltage levelmay undesirably change the voltage value of the down-converted signal.As a result, the actual voltage value of the down-converted signal maybe difficult to ascertain by down-stream processing.

[0417] Generally, “re-radiation” is an undesired phenomenon where asignal comprising one or more frequency components generated byreceiving circuitry is transmitted by an antenna. For example, thefrequency components may be generated by a local oscillator of thereceiving circuitry. When transmitted, these frequency components mayundesirably interfere with nearby receivers, or may be received back bythe same antenna that transmitted them. When the frequency componentsare received back by the same antenna that transmitted them, this may bereferred to “re-radiation recapture”. The phenomenon of re-radiationrecapture may further impair signals that are down-converted, and/or maycause undesirable DC offset voltages that may impair the down-convertedsignals. For instance, the re-radiated and recaptured signal may appearto the receiver as unwanted noise, within or without the frequencyband(s) of interest, or may combine with local signals to create anundesired DC offset voltage. The phenomenon of creating a DC offsetvoltage by re-radiation recapture is described further below. Solutionsprovided herein for eliminating unwanted DC offset voltages apply toeliminating DC offset voltages produced from re-radiation recapture.

[0418] Furthermore, signals in a receiver circuit may travel or radiateto other receiver circuit sections, causing problems similar to those ofre-radiation recapture described above, including problems of noise andDC offset voltages. For instance, local oscillator signals mayundesirably transmit through the circuit substrate, through the air, orthrough other paths, to other receiver circuit sections, causingunwanted noise problems and problems with unwanted DC offset voltagesbeing generated. Circuits provided herein for solving problems with DCoffsets, re-radiation, and re-radiation recapture also apply to solvingproblems of noise and unwanted DC offset voltages caused by thisphenomenon.

[0419] The concepts of DC offset and re-radiation are further describedin the following sub-sections. Furthermore, example methods and systemsare provided in subsequent sections below for reducing or eliminatingunwanted DC offset and re-radiation. Such methods and systems can beused alone, or in combination with each other, to address offset issues.

[0420] 7.1.1 Introduction

[0421] Embodiments of the UFT module may be used in many communicationsapplications. For some of these applications, the signal space mayinclude waveforms with near DC content. Such waveforms exist, forexample, in signals transmitted at radio frequencies. Hence, it may beadvantageous to limit the amount of artificial DC insertion or DCoffsets contributed by the UFT module or its complimentary demodulationarchitecture.

[0422] This section presents an overview of DC offset contributions ofthe UFT module, and related circuitry, relevant for zero IFimplementation. In addition, embodiments of the present invention arepresented for reducing the adverse impacts of the DC offsets.

[0423] 7.1.2 DC Offset Model Overview

[0424]FIG. 73 illustrates a down-conversion circuit 7300 according to anembodiment of the present invention. The down-conversion circuit 7300 ofFIG. 73 provides a model that indicates possible DC offsetcontributions. Down-conversion circuit 7300 comprises a UFD module 7302.UFD module 7302 comprises a UFT module (not shown).

[0425] There are at least three significant categories of offsets.

[0426] 1. Clock Excitation or Charge Injected

[0427] 2. Re-radiation Offsets

[0428] 3. Intermodulation Distortion

[0429] Each category possesses its own mechanisms.

[0430] The following definitions in Table 1 set the backdrop foranalysis and understanding of the offset phenomena from a high levelmodel. At least some of the phenomena relevant to the discussion interms of device physics may be lumped into one or more of the followingmodel parameters.

R(t)=[r(t)+k ₁ k ₂ C _(A)(t′ _(A))+k ₂ k _(B) C _(B)(t′ _(B)))+k _(LNA)+k _(A) C _(A)(t′ _(A))+k _(B) C _(B)(t′B)C(t)+<|C(t)|>k _(cλ)  Eq. 29

[0431] (*The charge injection path associated with k_(ff) has beenignored in Eq. 29. This component will be addressed separately in asubsequent section.) TABLE 1 r(t) = s(t) + r(t) is the received signalof interest which consists of a n(t);. modulated carrier s(t) and anoise component n(t). k₁k_(A)C(t′_(A1)) This signal is a conditionedclock 7304 or transient or waveform, which leaks to the core input ofUFD module k_(A)C(t′_(A)) 7302 across free-space, substrate, etc.t′_(A1) is a delayed time variable. t_(A1) = t − t_(A) − t₁ where t_(A)is the delay of the specific (A) path, and t₁ is the additional delaythrough the (1) path. k₂k_(B)C_(B)(t′_(B2)) This signal is similar tothe one described above except or that the leakage paths and delays aredifferent and the k_(B)C_(B)(t′_(B)) leakage signal is a raw clock 7306rather than a conditioned clock 7304. <|C(t)|>k_(cλ) This is a signalwhich is self-generating at UFD 7302 module, and is derived from thecharge injection phenomena at UFD 7302 module when the conditioned clock7304 or control is active. Essentially, the conditioned clock C(t) ismodified by a nonlinear operation (in this case an abs function)averaged or integrated over some interval and scaled by a gain constantk_(cλ) and delayed by t_(cλ). When C(t) is not active, then<|C(t)|>−k_(cλ) →0. This offset term is summed effectively at the outputof UFD module 7302. < >denotes the expectation operation. * k_(ff) Gainconstant associated with feed forward charge injection path. This pathis typically of interest when interferences are present. Usually,offsets will not be significant unless the S/I (Signal to Interferencepower) is very low and I is very large.

[0432] There may be additional leakage terms, which are not illustratedin the model.

[0433] 7.1.3 Clock Modulation via PN Code

[0434] A system and method for addressing DC offset, according to anembodiment of the present invention, involves modifying the LO (localoscillator) in such a manner that the offsets are randomized andspectrally spread. After some amount of amplification the randomizedsignal may be de-spread coherently. At least some of the offset,particularly that offset which is due to LO re-radiation, may beremoved. FIG. 74 illustrates a down-conversion circuit 7400, accordingto an embodiment of the present invention, that removes at least someoffset. Down-conversion circuit 7400 comprises a UFD module 7402. UFDmodule 7402 comprises a UFT module (not shown).

[0435] Although only a single down-conversion channel is illustrated inFIG. 74, in alternative embodiments the architecture may be extended toinclude both I and Q, especially if two uncorrelated PN (Pseudo-randomNoise) sequences are utilized. Other dual or multiple channelembodiments are also within the scope and spirit of the presentinvention. In an embodiment, the PN code or similar sequence is formedby a maximal length linear feed back shift register (or other logic) andis modulated onto the clock and pulse conditioned to form C(t).C_(PN)(t′) is the baseband PN sequence waveform. C_(PN)(t′) is virtuallyidentical to C_(PN)(t) except for a very small time shift. R′(t) may begiven by:

R′(t)=k _(BB) R(t−t _(BB))(C _(PN)(t′))  Eq. 30

[0436] Which may be expanded to: $\begin{matrix}\begin{matrix}{{R^{\prime}(t)}{k_{BB}\left\lbrack {{\left( {{{r\left( {t - t_{BB}} \right)}{C\left( t^{\prime} \right)}} + {\left( {t - t_{BB}} \right)}} \right)k_{LNA}} + {X\left( {t - t_{BB}} \right)}} \right\rbrack}} \\{{C_{PN}(t)} + {\left( {< {{\overset{\sim}{C}\left( {t - t_{BB}} \right)}} > {k_{c\quad \lambda}k_{BB}}} \right){C_{PN}\left( t^{\prime} \right)}}}\end{matrix} & {{Eq}.\quad 31}\end{matrix}$

[0437] where: $\begin{matrix}\begin{matrix}{{(t)} = {\left( {{k_{1}k_{A}{C\left( t_{1A}^{\prime} \right)}} + {k_{2}k_{B}{C_{B}\left( t_{2B}^{\prime} \right)}}} \right){C(t)}}} \\{{X(t)} = {\left( {{k_{A}{C\left( t_{A}^{\prime} \right)}} + {k_{B}{C_{B}\left( t_{B}^{\prime} \right)}}} \right){C(t)}}} \\{t_{x}^{\prime}{\underset{\_}{\Delta}\left( {t - t_{x}} \right)}} \\{{C(t)} \simeq {C_{A}\left( t^{\prime} \right)}}\end{matrix} & {{Eq}.\quad 32}\end{matrix}$

[0438] It will be apparent to persons skilled in the relevant art(s)from the teaching herein that examination of these equations, combinedwith the knowledge that C(t) can be a pseudo random sequence, willreveal interesting cross correlations for the math provided above.

[0439] For the moment, delays on the order of sub-carrier cycle timesand carrier cycle times may be ignored, thereby considering many of thedelay terms to be zero. While this may not actually be the case, thisdoes provide a substantially worst case bounding view ofcross-correlation properties of the described signal in one dimension.The general result would apply to the I/Q complex signal representation.However, the first step for a single dimension is instructive andtherefore provided. The cross-correlation R_(xx) is calculated asfollows:

R _(XX)(t)Δ<k _(BB) R(t)C _(PN)(t)>≈<k _(BB) R(t−t _(BB))C_(PN)(t′)>  Eq. 33

[0440] The result is:

R _(XX)(t)≅<[k _(BB) k ^(LNA) r(t)+k _(1A) k _(LNA) k _(BB) C(t)C_(A)(t)C _(PN)(t′)+k _(2B) k _(LNA) k _(BB) C(t)C _(B)(t)C _(PN)(t′)+k_(BB) k _(A) C _(A)(t′)C(t)+k _(BB) k _(B) C _(B)(t)C _(PN)(t′)+<|{tildeover (C)}(t)|>k _(cλ) k _(BB) C _(PN)(t′)[+  Eq. 34

[0441] C_(PN)(t), C_(A)(t), r(t), and C_(B)(t) average to zero over along term, if C_(PN)(T) is augmented. Even if r(t) does not average tozero, r(t) is not considered because it is the signal of interest.(NOTE: An “augmented” sequence refers to the process of chip stuffing asrequired to provide ideal code balance.)

[0442] It will be known to persons skilled in the relevant art(s) thatC(t) and C_(B)(t) are uncorrelated. It is also known that |C(t)| andC_(PN)(t) are uncorrelated when C_(PN)(t) is bipolar. If thecross-correlations indicated above are in fact indicative of theprocess, then R_(XX)(t) would approximately reduce to:

R _(XX)(t)≈→0  Eq. 35

[0443] R_(XX)(t) represents the DC offset that exists due to LOre-radiation leaking into the front end of UFD module 7402 by someancillary path, such that it is converted into band at the output of UFDmodule 7402 for the case where the leakage is synchronous in part orwhole to the UFD module transform, plus charge injected offset. Thissynchronicity is actually rare for cases where k_(1A) or k_(A) is large.Typically those gains would be much less than 1.

[0444] What the above equation reveals is that little or no DC offseteffects remain if C_(PN)(t) and C(t) are balanced, bipolar sequences.

[0445] In this case, a spreading sequence, spreading rate, and sequencelength are selected. This selection typically requires carefulexamination of the signaling scheme, data rate, etc. Also, the Cl pathinvolving k_(ff) has not been accounted for in this analysis.

[0446] 7.1.3.1 Interpretation of R_(xx)(t) and Required Leakage

[0447] It may be desirable that R_(xx)(t) be 3.16×10⁻⁶ volts peak in a50 Ω system for a number of applications. For a system design where theUFD module possesses an output impedance of 1 KΩ, a signal level of 63.2μV peak may be tolerated (−100 dBm).

[0448] In embodiments, clock port or control port signals may swing asmuch as 2V peak internal to the UFD module. If 2 volts must be reducedto 63.2 μV at the UFD module output, then: $\begin{matrix}{{{2{V \cdot k_{c\quad \lambda}}} < {63.2 \times 10^{- 6}V_{peak}}}\therefore{k_{c\quad \lambda} < \frac{63.2 \times 10^{- 6}}{2}}} & {{Eq}.\quad 36}\end{matrix}$

[0449] Hence:

k _(cλ)<31.6×10⁻⁶{tilde over (<)}−90 dB(power)  Eq. 37

[0450] Eq. 37 implies that the effective isolation from charge injectedDC must be on the order of 90 dB (power) or greater at the UFD module invarious embodiments.

[0451] In embodiments, it is unlikely that 90 dB of chip isolation wouldbe achieved in a system-on-a-chip design. It may be more difficult tomaintain isolations over temperature and production lots. In anembodiment, the suppression is such that the LO re-radiation inband@2450 MHZ for a n=5 system is −20 dBm.

[0452] A similar calculation for the aggregate LO re-radiationcomponents reveals the requirement of approximately 100 dB suppression,effectively.

[0453] 7.1.3.2 Charge Injected DC Offset

[0454] The charge injected DC offset phenomena may be modeled as somerectification of the clock or control port energy weighted by some gainconstant, k_(cλ). The amount of DC offset introduced at the output ofthe UFD module may be given as:

CI _(UFDDC)Δ(<|{tilde over (C)}(t)|>)k _(cλ)  Eq. 38

[0455] However, it may also be of value to construct a picture moreclosely associated with how this term arises. Consider a down-conversioncircuit 7500 shown in FIG. 75, configured according to an embodiment ofthe present invention. Down-conversion circuit 7500 comprises a UFDmodule 7502. UFD module 7502 comprises a UFT module (not shown). Anequation can be written to describe the voltage at the output due toC(t). The complex domain equation is: $\begin{matrix}{{V_{OCI}(s)} = {{V_{C}(s)}\left( \frac{{Z_{L} \cdot Z_{s}^{\prime}}{C_{OLeff} \cdot s}}{{\left. {{Z_{L}Z_{C}} + {Z_{S}^{\prime}Z_{C}} + {Z_{L} \cdot Z_{S}^{\prime}}} \right){C_{OLeff} \cdot S}} + \left( {Z_{L} + Z_{S}^{\prime}} \right.} \right)}} & {{Eq}.\quad 39}\end{matrix}$

L  Δ _     Complex     Load     Impedance  ( R L R L  C · S + 1)

Z s ′     Δ _     Complex     Source     Impedance  ( D2D + L ·S     S  ( s ) L     S + S  ( s ) )

[0456] V_(C)(s)Δ Complex Clock Signal driving the UFD module

[0457] V_(OCI)Δ The Complex Output Signal arising from clock activity atUFD module 7502. This component is considered as a voltage resultingfrom charge injection due to the parasitic C_(OLeff)

[0458] There are some high level considerations which reveal importantaspects of the phenomena. The equation shows the following;

[0459] When V_(C)(s) is a pure DC waveform, V_(OCI) is zero. However,V_(C)(t) does possess both a transient and DC offset component. If theDC offset component is zero then V_(C)(t) would also be zero.

[0460] As the frequency content of the transients in V_(C)(s) are lower,then so too V_(OCI) will typically become lower. However, this perceivedmonotonic correspondence of V_(OCI) to frequency components V_(C)(s) maynot always hold because of resonances in the complex impedancessurrounding UFD module 7502. The DC offset performance of UFD module7502 is a strong function of the Fourier signature for V_(C)(t), as isfurther described below.

[0461] When C_(OLeff)→zero, then V_(OCI)→zero.

[0462] The lower the source impedance and the lower the load impedance,the lower V_(OCI) becomes.

[0463]_(UFD module) tends to provide some isolation from inputimpedances over the frequency ranges where the real [_(UFD module)]series component dominates. When real [_(UFD module)] is significant,_(L) becomes a consideration concerning V_(OCI).

[0464] C_(OLeff) is a parasitic which is well known and understood inconventional receiver systems. There are processes available which canreduce this parameter by a factor approaching 100. A value in oneembodiment of UFD module 7502 would be on the order of:

C _(OLeff)≅120 pf  Eq. 40

[0465] Hence, this could be reduced to 1-2 pf.

[0466] The amount of charge injected DC voltage variation at the outputof UFD module 7502 is related to one or more of at least the followingfactors:

_(S)(s): The Input Source Impedance, which is typically complex. L: Thisis a typically used bias inductor. Other arrangements are possible. Thisone is selected simply for illustration purposes.

_(C)(s): The Output Impedance of the Clock Source (C(t)). C,R_(L):Components utilized to load UFD module 7502. C_(OLeff): This capacitanceis a process parasitic and is shown as an effective capacitor formedfrom several physical capacitors, which usually dominates in terms ofcharge injection path. Although shown on the output node it may beactually split between output and input of UFD module 7502. In fact, theinput typically provides a significant LO re-radiation path.

[0467]_(UFD module): Internal Impedance of UFD module 7502.

[0468] Because ′_(S), _(L), and _(C) are all complex impedances, thereis always the chance that resonance's may occur for certain C_(OLeff)such that V_(OCI) could possess local maxima even as C_(OLeff)decreases. In the case where _(UFD module), _(L), ′_(S), and _(C) aredominated by real parts, the injection attenuation gains in droppingC_(OLeff) from 120 pf to 2 pf are enormous. These attenuation gains maybe roughly 35 dB in power, and half that in voltage. Hence, DC offsetdue to charge injection may be significantly attenuated by processcontrol. An example of process control may be related to moving fromCMOS (Complementary Metal Oxide Semiconductor) to DMOS (Double DiffusedMetal Oxide Semiconductor). There are processes available which mayinclude both CMOS and DMOS on the same substrate, possibly providingimportant performance options, particularly in the domain of gateoverlap capacitance control. The effective gate overlap capacitanceC_(OLeff) is a chief offender, which results from process oxidecapacitance in conjunction with overlap parasitics related to transistorgeometries.

[0469] Another method of artificially decreasing C_(OLeff) is bychanging _(C) to incorporate a series capacitor, which is much lowerthan C_(OLeff). However, this must be done carefully to avoid negativesubstrate transients. A further useful circuit model allows ′  s -> 0 ,C -> 0 , D2D -> 0 , Z L -> 1 sC .

[0470] An embodiment of this circuit is shown in FIG. 76, asdown-conversion circuit 7600. Down-conversion circuit 7600 comprises aUFD module 7602. UFD module 7602 comprises a UFT module (not shown).Under these conditions: $\begin{matrix}{V_{OCI} \approx {{- \left( \frac{C_{OX}}{2C} \right)}\left( {W \cdot L} \right)\left( {V_{cp} - V_{T} - V_{i\quad n}} \right)}} & {{Eq}.\quad 41}\end{matrix}$

[0471] where: C_(OX) Δ Oxide Capacitance, Function of Process. W, LΔFundamental Geometries related to the UFD module, which affect parasiticoverlap capacitances. V_(cp) Δ Conditioned Clock Peak Excursion(unfiltered). V_(T) Δ Threshold Voltage related to the Process.

[0472] Eq. 41 relates directly to the device physics of the UFD module.C_(OLeff) relates to C_(OX) and the parasitics formed due to W and L.

[0473] This model has some practical application because it can be usedto predict compromises in the charge injection DC offset due to UFDmodule 7602 process parameters and the output capacitor C. For example,the model can predict, to a reasonable approximation, the results of acorresponding simulation. In the situations where _(S), _(C), and_(UFD module) may not be precisely known, a circuit designer may atleast select approximate specifications for UFD module 7602 designsusing the simple model, and add more accurate impedances as they becomeknown. Furthermore, to the degree C_(OX), W, L, V_(T), and V_(cp) can bemanipulated, the more V_(OCI) can be reduced.

[0474] 7.1.3.3 Clock Waveform Impact on CI Induced Offsets

[0475] The previous section illustrated that the clock waveform canimpact the efficiency of CI DC offset build up. This is an importantconcept because clock design is integral to the UFD module theory. Thefollowing formulation provides a Fourier series representation for ageneral clock pulse, and provides some insight into the frequencycontent of the excitation clock. The DC introduced by charge injectionis a strong function of complex impedances around the UFD module.Signals which stimulate the UFD module may also play a role in the DCoffset, depending on the clock signal's Fourier signature.

[0476] Clock pulse train V_(C)(t) may be represented by a Fourier seriesas follows: $\begin{matrix}{{V_{C}(t)} = {\frac{V_{cp}T_{A}}{4} + {\sum\limits_{n = 1}^{\infty}\quad {2{V_{cp}\left( \frac{\sin \left( {n\quad \pi \quad {T_{A}/2}} \right)}{n\quad \pi} \right)}\left( {\cos \left( \frac{2\quad n\quad \pi \quad t}{T_{S}} \right)} \right)}}}} & {{Eq}.\quad 42}\end{matrix}$

[0477] An ideal rectangular clock pulse is illustrated in FIG. 77A, withzero rise and fall time. This may be considered to be a worst casescenario. In reality the clock waveform will consist of a repeatingpulse train with a basic pulse shape possessing finite, non-zero, rise,and fall times.

[0478] The calculated Fourier series for V_(C)(t) is a well known resultfor sampling devices. As T_(A) decreases, the Fourier spectrum extendsever greater in the frequency domain with significant harmonics.

[0479]FIG. 77B illustrates that the spectrum is a “picket fence” withharmonics separated by f_(S)=T_(S) ⁻¹, and nulls at N·f_(a), wheref_(a)=T_(A) ⁻¹. Hence, the greater the value of ratio T_(S)/T_(A), thegreater the number of harmonics out to the first null, and the greaternumber of components, spectrally, which can excite the process parasiticat higher and higher frequencies.

[0480] A sequence of plots in FIGS. 79-83 and 86A illustrate thisconcept of clock waveform attributes and relationship to the DC offset.

[0481]FIG. 78 illustrates a down-conversion circuit 7800 used todetermine DC offset due to charge injection, and possibly LO feedthrough, according to an embodiment of the present invention.Down-conversion circuit 7800 comprises a UFD module 7802. UFD module7802 comprises a UFT module (not shown).

[0482]FIG. 79 shows the offset obtained with 3 different clock pulsewidths (TA) for the circuit of FIG. 78. The clock is configured tooperate on a 2.4-2.5 GHz band signal using a 5th harmonic technique. Theclock for the offset plots of FIG. 79 was selected to down-convert 2.45GHz.

[0483]FIG. 79 shows that longer pulse widths may produce lower CIrelated offsets. FIG. 80 illustrates a situation similar to that of FIG.79 utilizing a 3rd harmonic clock approach. As shown in FIG. 80,reducing the clock frequency components did not continue to reduceV_(OCI). This may be due in part to the surrounding complex impedanceswhich will possess local resonances or favor certain Fourier spectrums.

[0484] The circuitry surrounding UFD module 7802 may affect overallcircuit performance. For example, FIG. 81 shows offsets obtained usingslightly lower bond wire inductance. FIG. 81 illustrates how the resultsof FIG. 79 may be affected by these changes.

[0485]FIG. 82 illustrates the case of a fixed 204 ps T_(A) with 10 psrise and fall times, while permitting a variation in bond wireinductance. FIG. 82 indicates that lower inductance may be better insome situations.

[0486] The previous plots related to cases with clock waveforms V_(C)(t)possessing 10 ps rise and fall times. FIG. 83 illustrates the V_(OCI)response for a variety of rise and fall times, 204 ps T_(A), and 5thharmonic operation.

[0487] It is interesting to note that there are two local minima for theDC offset performance with fast rise times representing one of thosecases. This implies resonance in the complex impedances surrounding (andincluding) UFD module 7802. Different circuit topologies will behavedifferently and different component types would operate differently dueto their own parasitic elements. In addition, stretching to a 3× or 5×aperture would produce different results.

[0488] 7.1.3.4 Bench Example

[0489] Experiments were conducted with hardware designed to operate inthe 2.4 GHz ISM (Industry, Scientific, and Medical) band. 5th harmonicmode was utilized for the clock, with the clock rate being variedbetween 482.4 MHZ and 492.4 MHZ. The UFD module configuration was an I/Qreceiver with matching networks and DC coupling. The input to the I/Qassembly was terminated with 50 Ω.

[0490] The charts shown in FIGS. 84A, 84B, 85A, and 85B record theresults on the I port for a variety of LO drive levels and 3 operatingchannels, for two different assemblies: one with a clock port match andone without.

[0491] 7.1.3.5 Complementary Architecture

[0492] Up to this point the UFD module cores analyzed have been based ona non-complimentary structure. Complementary structures can be used withthe important advantage of lower UFD module losses and greater IP2, IP3performance. In addition, some charge injection cancellation should bepossible. The results in FIG. 86A correspond to results recorded in FIG.79. Careful examination shows that there may be a 4.25 dB reduction inCI induced DC offset possibly attributed either to the complementary UFDmodule architecture or the resulting modification to _(UFD module).

[0493] 7.1.3.6 Spreading Code Results

[0494] Sections 7.1.3 and 7.1.3.1 outline the concept of using a localPN code to reduce the DC offset generated at a UFD module, or createddue to LO re-radiation recapture. Maximal length codes, balanced codes,and other related code types may be used. Furthermore, the statisticalproperties of a code may be tailored in the time domain or frequencydomain to accomplish desired DC reduction while minimizing the impact tothe desired signal.

[0495] Experiments have been accomplished with UFD module circuitembodiments to illustrate the potential of these techniques. FIG. 86Bshows an example spectral plot of a carrier tone at RF, corresponding toLO re-radiation at a UFD module.

[0496]FIG. 86C illustrates the LO re-radiation spectrum shown in FIG.86B after modulation by an example modified maximal length linear PNsequence. In this example, the power spectral density is substantiallymodified by the code. On average, the power spectral density has beenlowered by approximately 32 dB. This benefit may not be completelyobtained unless the specification desired for re-radiation is referencedto the resolution bandwidth of the analyzer sweep. This is adjusted forwider bandwidth (faster) PN sequences with long repetition intervals.That is, the processing gain of interest here is the bandwidth expansionfactor: $\begin{matrix}{{BW}_{E} = \frac{{BW}_{pn}}{{resBW}_{spec}}} & {{Eq}.\quad 43}\end{matrix}$

[0497] Where: BW_(E) Δ Bandwidth expansion factor (unitless) BW_(pm) ΔDouble Sided Bandwidth of PN Sequence resBW_(spec) Δ Resolutionbandwidth for a particular re-radiation specification that is dictatedby standards or regulatory agency.

[0498] Then:

P _(BW)=10log₁₀(BW _(E))dB  Eq. 44

[0499] P_(BW) is the effective processing gain due to LO bandwidthexpansion factor alone, that is attained by using a special sequence ata UFD module clock port superposed on the clock. To some extent, BW_(pm)can be adjusted for a desired effect, although there may be otherpractical system constraints.

[0500] As predicted by equations in section 7.1.3, the DC offset at aUFD module output may be canceled using a special sequence, itscorrelation properties, and its effective system processing gain.

[0501]FIG. 86D shows an example PN modulated output of a UFD moduleconfigured to receive a 870 MHZ RF signal with a slight carrierfrequency offset. A beat note represents the slight carrier offset(envelope of the baseband). In addition, the PN code impressed on thereceived signal by the special UFD module clock signal is visible.

[0502] The signal illustrated in FIG. 86D possesses substantial DCoffset. FIG. 86E illustrates the result after PN rectification orcorrelation. The DC offset produces a PN code summed to the desiredsignal while the balanced PN modulation envelope is removed bycorrelation. The power in the remaining summed PN signal is directlyproportional to the original UFD module DC offset plus all systemoffsets thereafter up to the post-correlator. The bandwidth of thisancillary PN code power is substantially wider than the bandwidth of thebaseband signal by design. Hence, the post filter (sometimes a basebandmatched filter) can remove much of the variance of the PN sequence. FIG.86F illustrates the low pass output to recover the baseband beat note.

[0503] A goal is to choose an effective system processing gain PG_(sys),which is high enough to drive significant variance from the low passresult. PG_(sys) is defined as follows: $\begin{matrix}{{PG}_{sys}\underset{\_}{\Delta}\quad 10\log_{10}\frac{\left( {{BW}_{pn}/2} \right)}{{BW}_{MF}}} & {{Eq}.\quad 45}\end{matrix}$

[0504] BW_(pm)Δ A Double Sided PN Code Bandwidth

[0505] BW_(MF)Δ BB Filter Bandwidth

[0506] The example run in the lab utilized a 10 kHz baseband signalbandwidth and a spreading rate of 5 MHz. In addition, the code wasmodified as an R type. This technique may not provide all of the DCcancellation required but can be a powerful tool for many applications.

[0507] 7.1.4 UFD Module DC Offsets From Non-Linearities

[0508] Because the UFD module is at least a conversion device, anintercept point will determine output waveform integrity to a largeextent. Two tone 2^(nd) order intercept and two tone 3^(rd) orderintercept points are important. In particular, the two tone second orderintercept point, IP2_(IN), relates to DC offset. As the input begins toapproach the UFD module rails, harmonic spectrums are generated in thesignal path. Because the UFD module clock may excite harmonics, eachharmonic spectrum may down-convert to DC, adding some DC offset. Becausethe phases of the down-conversion harmonics generally are complicated,the resulting DC offset may be non-systematic, even though the processis predictable by using complex math.

[0509]FIG. 86G illustrates an exemplary signal input harmonic spectrumand conversion clock harmonic spectrum. The harmonic spectrums for theinput signal at f₂ and f₃ become more significant as the UFD module ispushed harder on its input.

[0510] Another concept useful in considering the IP2_(IN) mechanismcomes from a different view on the frequency doubling phenomena.Frequency doubling occurs in a square law device. Hence, for the 2^(nd)order term, the non-linearity from the UFD module output may beapproximated by;

(Ã(t)cos(ω+t+φ(t)))²=½Ã(t)₂(1+cos(2ω₀ t+2φ(t)))  Eq. 46

[0511] Ã(t)Δ Amplitude Domain Modulation

[0512] φ(t)Δ Phase Domain Modulation

[0513] Ã(t) could represent the complex envelope of modulation frominformation impressed on the carrier (as well as noise). Likewise, φ(t)could contain information modulated onto the carrier as well as phasenoise. The above equation illustrates that the 2× frequency componentcan be formed from the non-linearity but that is Ã(t)² is also formed.The equation indicates that a DC component results from the squaredenvelope. This DC component is not desirable. Likewise, higher orderinter-modulation can contribute to the problem, particularly even orderterms.

[0514] In general, the output voltage of a non-linear system can beexpanded in terms of its input voltage by a power series of the formshown in FIG. 86H. Usually it is difficult to predict k₁, k₂ . . .precisely. Extending properties of linear systems to non-linear systemdescriptions permits another useful and more general equation:

y _(n)(t)=∫_(∞) ^(∞) . . . ∫k _(n)(u ₁ , u ₂ , . . . u _(n))X(t−u₁)X(t−u ₂) . . . X(t−u _(n) du ₁ , du ₂ . . . du _(n))  Eq. 47

[0515] where y_(n)(t) is the system output and X(t) is the system input.This is the so called nth order impulse response for the system, foundby an n-fold convolution kernel. FIG. 86I shows a block diagramrepresentation of this system.

[0516] y₁(t) is the desired linear impulse response of the system. y₂(t)is the two-dimensional system convolution involving X(t). y₃(t) is thethree-dimensional convolution of X(t) and the impulse response h₃(u₁,u₂,u₃), etc. This is known as the Volterra functional seriesrepresentation of a system. For weak non-linearities, the first 3 termsof the series may provide enough information to characterize a system.This is the case for many communications systems.

[0517] Such nth order analysis in practice is often complex and tedious,yielding only a general feel for the expected result, unless circuit andnetwork models are extraordinarily accurate. Nevertheless, in theapproximation, the 2^(nd) order term relating to the two tone 2^(nd)order input intercept (IP2_(IN)) is one useful metric for measuringdown-conversion linearity. Essentially, the DC offset from IP2_(IN) isbounded at the upper end by the power of the 2^(nd) order harmonic.

[0518] For instance, suppose that it is desired to suppress the power ofthe 2^(nd) order term by 20 dB in a direct down-conversion device (nointerference present). If the highest expected input RF signal ofinterest is −25 dBm, the system will require an input intercept(IP2_(IN)) of −5 dBm. This establishes a signal-to-DC offset ratio of atleast 20 dB due to the 2nd order non linearity.

[0519] Now consider the case where other unwanted signals are present atthe input to the non-linearity along with the signal of interest.Suppose the RF signal is at a level of −101 dBm, while the interferencetone is a level of −30 dBm. Furthermore, assume that the system noisefloor is near enough to −101 under linear conditions such that we desirean additional 10 dB margin on any 2^(nd) order non-linearity folded backin band, so that our benchmark at 101 dBm is not affected.

IP2_(IN)=(−30+111)−30=51 dB  Eq. 48

[0520] Therefore, IP2_(IN) can become a significant specification whenan interference or blocking tone is considered, and unfiltered due to azero IF architecture.

[0521] This type of non-linear effect is dependent on input signal powerto a great extent. Because the phenomenon is based on even-orderintermods, differential design can cancel a significant portion of thedifficulty, but imbalance may not remove it all. For instance, supposethat the input to the UFD module is at −15 dBm due to an LNA in front ofthe down-conversion. Suppose this is an interfering tone. Also, assume a1 KΩ baseband operating impedance and a 12 dB UFD module conversionloss. The suppression of the IP2_(IN) at the UFD module output is then:

DC(IP ₂)≦−15−12−8  Eq. 49

[0522] If the desired suppression is 81 dB, the output offset into 1 KΩis less than 0.126 mV due to 2^(nd) order non linearities. This may beaccommodated with an op amp circuit, for example. If a differentialarchitecture is assumed, then arguably this signal can be processed interms of common mode range.

[0523] In fact, in embodiments a UFD module with IP2_(IN) of +40 dBmcould be tolerated if 10 dB of cancellation is available from adifferential architecture. Differential architectures may extend asgreat as a 30 dB benefit, for example, without special trimming.

[0524] 7.2 Example Embodiments to Address DC Offset and Re-RadiationProblems

[0525] Section 7.1 above discussed problems related to DC offset andre-radiation that occur during and after the down-conversion process,and were provided for illustrative purposes, and are not limiting.Embodiments were also provided for reducing or eliminating unwanted DCoffset and re-radiation using techniques of spectral spreading followedby de-spreading, according to the present invention. Various embodimentsrelated to the problems, method(s), and structure(s) described above arepresented in this section (and its subsections). In particular, furtherapplications of the UFT module are provided below in circuitconfigurations that reduce or eliminate problems of DC offset andre-radiation.

[0526] These embodiments are described herein for purposes ofillustration, and not limitation. The invention is not limited to theseembodiments. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. The invention is intended and adapted toinclude such alternate embodiments.

[0527] 7.2.1 DC Offset

[0528] Exemplary embodiments are provided below for reducing oreliminating unwanted DC offset voltages. These unwanted DC offsetvoltages include unwanted DC offset voltages created by any source,including non-ideal circuit component operation, re-radiation recapture,local circuit signals traveling or radiating to other circuit sections,etc. The embodiments provided below are not limited to this use, but mayhave additional applications. For example, these embodiments may beapplicable to reducing or eliminating unwanted circuit re-radiation.

[0529] 7.2.1.1 Reducing DC Offset by Spectral Spreading and De-Spreading

[0530] Embodiments for reducing DC offset by spectral spreading andde-spreading, as described above, are further described in the followingsub-sections, and additional related embodiments are presented.

[0531] 7.2.1.1.1 Conventional Wireless Communications Receiver

[0532]FIG. 87 shows an example conventional wireless communicationsdown-conversion system 8700. Down-conversion system 8700 comprises adown-conversion module 8702 and an amplifier 8704. Down-conversionmodule 8702 typically comprises a super-heterodyne receiver.Down-conversion module 8702 may comprise multiple down-conversionstages. Amplifier 8704 may comprise an amplifier, a filter, other signalprocessing component(s), or any combination thereof.

[0533] Down-conversion module 8702 down-converts a modulated carriersignal 8706, according to at least one local oscillator signal 8708, toa down-converted baseband signal 8710.

[0534] Down-converted baseband signal 8710 is input to amplifier 8704.Amplifier 8704 amplifies, filters, and/or otherwise processesdown-converted signal 8710, and outputs baseband signal 8712.

[0535] As described above, and shown in FIG. 87, DC offsets due to localoscillator signal 8708 may be input to the signal path at severalpoints, with some possible points indicated by charge leakage and chargeinjection paths 8714, 8716, and 8718. As described above, charge leakageand charge injection are well known effects. These DC offsetsdisadvantageously affect at least the dynamic range and accuracy ofbaseband signal 8712. For instance, adding a significant DC offset tobaseband signal 8712 may cause the output of subsequent amplifiers inthe baseband signal path to approach the level of their power supplies,potentially causing the amplifiers to rail or become non-linear.

[0536] 7.2.1.1.2 Spread/De-Spread Receiver Embodiments of the PresentInvention

[0537] An embodiment of the present invention addresses undesired DCoffsets described above by modifying the local oscillator in such amanner that offsets are randomized and spectrally spread. Thispseudo-random local oscillator signal is used to down-convert an inputsignal, such as a modulated carrier signal, and spread the spectrum ofthe down-converted signal. After some amount of amplification,filtering, and/or other optional processing, the randomizeddown-converted signal may be spectrally de-spread to a baseband signal.Because the down-converted signal is spectrally de-spread, offsets arespectrally spread. At least some of the offset, particularly the offsetdue to local oscillator re-radiation, is reduced or removed from theresulting baseband signal. The offset is spread over a frequency range.

[0538]FIG. 88A shows an exemplary spreader/de-spreader down-conversionsystem 8800, according to an embodiment of the present invention.Spreader/de-spreader down-converter system 8800 comprises a UFD module8802, an amplifier 8804, a first multiplier 8806, a second multiplier8808, an oscillator 8826, a pulse shaping circuit 8828, and a codegenerator 8832. UFD module 8802 comprises at least one UFT module.Amplifier 8804 may introduce an unwanted DC offset voltage onto a signalbeing down-converted by system 8800. Spreader/de-spreaderdown-conversion system 8800 operates to reduce or eliminate thisunwanted DC offset voltage.

[0539] Oscillator 8826 outputs oscillating signal 8830. FIG. 88C showsan example waveform for oscillating signal 8830. Oscillating signal 8830is preferably a periodic sine wave, but may be other periodic signalwaveforms such as square wave, triangle wave, ramp wave, and otherwaveforms.

[0540] Code generator 8832 outputs coded sequence signal 8816. Codedsequence signal 8816 is preferably a signal coded according to apseudo-random code sequence. For example, acceptable pseudo-randomcoding includes PN coding. Other applicable code schemes such as arealso within the scope of the present invention, such as square waves andManchester encoding. FIG. 88D shows at least a portion of an examplecoded sequence signal 8816.

[0541] First multiplier 8806 receives oscillating signal 8830 and codedsequence signal 8816. First multiplier 8806 multiplies oscillatingsignal 8830 and coded sequence signal 8816, and outputs a codedoscillating signal 8814. Coded oscillating signal 8814 comprises atleast some cycles of oscillating signal 8830 modified (or spread orcoded) according to coded sequence signal 8816. FIG. 88E shows anexample waveform for coded oscillating signal 8814.

[0542] In a preferred embodiment, when coded sequence signal 8816 is a“high” signal and/or represents a “1”, the phase of correspondingcycle(s) of oscillating signal 8830 are not modified, and when codedsequence signal 8816 is a “low” signal and/or represents a “0” or a“−1”, the phase of corresponding cycle(s) of coded oscillating signal8814 are shifted 180 degrees. For example, as shown in FIG. 88H, in thetime that occurs prior to time line 8834, coded sequence signal 8816 ishigh, and hence coded oscillating signal 8814 is essentially equal tooscillating signal 8830. In the time occurring between time lines 8834and 8836, coded sequence signal 8816 is low, and hence coded oscillatingsignal 8814 is essentially equal to oscillating signal 8830 with itsphase shifted by 180 degrees.

[0543] Pulse-shaping circuit 8828 inputs coded oscillating signal 8814.The output of pulse-shaping circuit 8828 is a coded control signal 8818,which preferably comprises a string of pulses. Coded control signal 8818comprises at least some pulses that are modified (or spread or coded)according to coded sequence signal 8816. FIG. 88F shows an examplewaveform for coded control signal 8818. Pulse-shaping circuit 8828controls the pulse width of pulses of coded control signal 8818.

[0544] UFD module 8802 receives an input RF signal 8812 (although itcould be an unmodulated signal) and coded control signal 8818. FIG. 88Bshows an example waveform for input RF signal 8812. UFD module 8802frequency down-converts and spectrally spreads input RF signal 8812 todown-converted spread spectrum signal 8820, according to coded controlsignal 8818. FIG. 88G shows an example waveform for down-convertedspread spectrum signal 8820.

[0545] For example, FIG. 88F shows an embodiment where coded controlsignal 8818 is PN coded. For a positive PN code chip (for example, priorto time line 8834), the input RF signal 8812 is effectivelydown-converted to down-converted spread spectrums signal 8820 in anormal, non-inverted fashion. For a negative PN code chip (for example,between time lines 8834 and 8836), the phase of one or more cycles ofcoded control signal 8818 are shifted by 180 degrees, and therefore theopposite phase of input RF signal 8812 is sampled. Hence, for a negativePN code chip, a segment of input RF signal 8812 is effectively invertedand down-converted to down-converted spread spectrum signal 8820.

[0546] Down-converted spread spectrum signal 8820 is optionallyamplified and/or otherwise processed by amplifier 8804 (or othercircuitry or processing modules), and a processed down-converted spreadspectrum signal 8822 results.

[0547] Unwanted DC offset may be summed into down-converted spreadspectrum signal 8820 during and after down-conversion and spectralspreading, and during and after processing by amplifier 8804.

[0548] Second multiplier 8808 receives coded sequence signal 8816 andprocessed down-converted spread spectrum signal 8822. Second multiplier8808 multiplies coded sequence signal 8816 and amplified down-convertedspread spectrum signal 8822. Down-converted spread spectrum signal 8822is spectrally de-spread in second multiplier 8808, and baseband signal8824 is output. FIG. 88H shows an example waveform for baseband signal8824. The unwanted DC offset is spectrally spread in second multiplier8808, reducing or removing the offset from baseband signal 8822.Baseband signal 8822 may be a baseband information signal, or may be anintermediate frequency (IF) signal.

[0549] For example, in an embodiment using PN coding, for a positive PNcode chip, amplified down-converted spread spectrum signal 8822 ismultiplied by 1 (not inverted) in second multiplier 8808. For a negativePN code chip, the amplified down-converted spread spectrum signal 8822is multiplied by −1 (inverted). In this manner down-converted spreadspectrum signal 8822 is spectrally de-spread.

[0550]FIG. 88H shows a pulse 8838 in the example waveform of basebandsignal 8824. Pulse 8838 may result from the phase shift of codedoscillating signal 8814 in multiplier 8806 causing a delay between edgesof coded sequence signal 8816 and down-converted spread spectrum signal8820. In preferred embodiments, each chip or pulse of control signal8816 may be equal in length to a substantial number of cycles ofoscillating signal 8830, potentially in the hundreds or greater (theexample of FIG. 88H does not show this). Because of this, pulse 8838will occur relatively infrequently on baseband signal 8824, is of highfrequency relative to baseband signal 8824, and hence may be filteredout of baseband signal 8824 by a conventional filter.

[0551] Although only a single down-conversion channel is illustrated inthe example embodiment of FIG. 88A, the present invention may beextended to two or more channel embodiments, including I/Q modulationsystem embodiments. The spreading sequence, spreading rate, and sequencelength may be selected according to the signaling scheme, data rate, andother factors, as would be apparent to persons skilled in the relevantart(s) from the teachings contained herein. Furthermore, the presentinvention is applicable to conventional down-converter embodiments, suchas shown in FIG. 89.

[0552]FIG. 108 depicts a flowchart 10800 that illustrates operationalsteps corresponding to the structures of FIGS. 88A and 89, fordown-converting and spectrally spreading an input signal, according toan embodiment of the present invention. The invention is not limited tothis operational description. Rather, it will be apparent to personsskilled in the relevant art(s) from the teachings herein that otheroperational control flows are within the scope and spirit of the presentinvention. In the following discussion, the steps in FIG. 108 will bedescribed.

[0553] In step 10802, an input signal is down-converted. In embodiments,the input signal is down-converted with a universal frequencydown-conversion module according to a coded control signal.

[0554] In step 10804, the down-converted input signal is spectrallyspread to a down-converted spread spectrum signal. In embodiments, step10804 may be at least partially integral with step 10802.

[0555] In step 10806, the down-converted spread spectrum signal isprocessed. For instance, the down-converted spread spectrum signal maybe amplified, filtered, or otherwise processed, as further describedabove. Furthermore, a DC offset voltage may be summed with thedown-converted spread spectrum signal, as described further above.

[0556] In step 10808, the down-converted spread spectrum signal isspectrally de-spread to a baseband signal. The down-converted spreadspectrum signal is multiplied with a code used to code the controlsignal. Furthermore, during this step, the DC offset voltage isspectrally spread, as further described above.

[0557] For illustrative purposes, the operation of the invention isoften represented by flowcharts, such as flowchart 10800 in FIG. 108. Itshould be understood, however, that the use of flowcharts is forillustrative purposes only, and is not limiting. For example, theinvention is not limited to the operational embodiment(s) represented bythe flowcharts. Instead, alternative operational embodiments will beapparent to persons skilled in the relevant art(s) based on thediscussion contained herein. Also, the use of flowcharts should not beinterpreted as limiting the invention to discrete or digital operation.In practice, as will be appreciated by persons skilled in the relevantart(s) based on the herein discussion, the invention can be achieved viadiscrete or continuous operation, or a combination thereof. Further, theflow of control represented by the flowcharts is provided forillustrative purposes only. Steps may occur in a different order thanshown. Furthermore, as will be appreciated by persons skilled in therelevant art(s), other operational control flows are within the scopeand spirit of the present invention.

[0558] 7.2.1.3 Charge Injection Reduction Embodiment

[0559] The spectral spreading/de-spreading embodiments described abovereduce or eliminate DC offset from a variety of sources. In thissection, an alternative embodiment, according to the present invention,is provided for reducing or eliminating DC offset due at least to chargeinjection. FIG. 90 illustrates some aspects of charge injection relatedto the present invention. FIG. 90 shows a UFD module 9000 comprising aUFT module 9002, a storage device 9004, and a reference potential 9006.In an embodiment, UFT module 9002 comprises a MOSFET 9008, and storagedevice 9004 comprises a capacitor 9010, although the invention is notlimited to this example.

[0560] An input RF signal 9014 is received by a first terminal 9028 ofMOSFET 9008. A control signal 9018 is received by a second terminal 9030of MOSFET 9008. A third terminal 9032 of MOSFET 9008 is coupled to afirst terminal 9034 of storage device 9004. A second terminal 9036 ofstorage device 9004 is coupled to reference potential 9006 such as aground 9012, or some other potential. In an embodiment, MOSFET 9008contained within UFT module 9002 opens and closes as a function ofcontrol signal 9018. As a result of the opening and closing of thisswitch, a down-converted signal, referred to as output signal 9016,results.

[0561] A well known phenomenon called charge injection may occur in sucha switching environment. As control signal 9018 applies a pulse waveformto the gate of MOSFET 9008, MOSFET 9008 is caused to open and close.During this operation, charge allowed to flow along a DC path 9024 maybuild on the gate-to-drain and/or gate-to-source junctions of MOSFET9008, as indicated on FIG. 90 as charge buildup 9020 (note that thesource and drain terminals of MOSFET 9008 are essentiallyinterchangeable). Charge buildup 9020 may leak from MOSFET 9020 throughleakage path 9022, and become stored on capacitor 9010. This charge thatbecomes stored on capacitor 9010 may cause a change in the voltageacross capacitor 9010. This voltage change may accordingly appear onoutput signal 9016 as a potentially non-negligible DC offset voltage.This non-negligible DC offset voltage on output signal 9016 may lead todifficulties in recovering the baseband information content of outputsignal 9016. Hence, it would be advantageous to reduce or prevent thispotential generation of DC offset voltage caused by this interaction ofcontrol signal 9018 with UFD module 9000.

[0562]FIG. 91 illustrates an exemplary circuit configuration forreducing unwanted DC offset voltage caused by charge injection,according to an embodiment of the present invention.

[0563]FIG. 91 shows UFD module 9000 of FIG. 90, with a capacitor 9126coupled between input RF signal 9014 and UFD module 9000. Capacitor 9126is preferably a small valued capacitor, such as, but not limited to, 10pF. The value for capacitor 9126 will vary depending upon theapplication, and accordingly its characteristics are implementation andapplication specific. Capacitor 9126 prevents DC current from flowingalong the path shown as DC path 9024 in FIG. 90, and thus reduces orprevents the flow of charge to, and build up of charge on capacitor9010. This in turn reduces or prevents a DC offset voltage resultingfrom the above described charge injection from appearing on outputsignal 9016. Hence, the baseband information content of output signal9016 may be more accurately ascertained.

[0564]FIG. 109 depicts a flowchart 10900 that illustrates operationalsteps corresponding to FIG. 91, for down-converting an input signal andreducing a DC offset voltage, according to an embodiment of the presentinvention. The invention is not limited to this operational description.Rather, it will be apparent to persons skilled in the relevant art(s)from the teachings herein that other operational control flows arewithin the scope and spirit of the present invention. In the followingdiscussion, the steps in FIG. 109 will be described.

[0565] In step 10902, an input signal is coupled by a series capacitorto an input of a universal frequency down-conversion module.

[0566] In step 10904, the input signal is frequency down-converted withthe universal frequency down-conversion module to a down-convertedsignal. The input signal is down-converted according to a controlsignal. The control signal under-samples the input signal.

[0567] In step 10906, a DC offset voltage in the down-converted signalgenerated during step 10904 is reduced. In an embodiment, the DC offsetvoltage is generated at least by charge injection effects due tointeraction of the control signal with the universal frequencydown-conversion module, as further described above.

[0568] It should be understood that the above examples are provided forillustrative purposes only. The invention is not limited to thisembodiment. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. The invention is intended and adapted toinclude such alternate embodiments.

[0569] 7.2.1.4Auto-Zero Compensation

[0570] Unwanted DC offset may be injected by circuit components in theintermediate frequency (IF) processing path or baseband processing pathfollowing a UFD module. In some cases, this DC offset voltage must bereduced or eliminated. In some situations, the output signaldown-converted by the UFD module may be a low level signal, where evensmall DC offsets inserted by components following the UFD module mayundesirably affect its value.

[0571] The previous section described inserting a series capacitor priorto the UFD module to reduce DC offset voltages due to charge injection.In embodiments, a capacitor may be added in series in the basebandprocessing path after a UFD module to reduce or eliminate DC offsetvoltages. In some situations, however, adding a capacitor in series inthe baseband processing path after the UFD module is not desirable. Forinstance, in some situations, it may be difficult to charge such aseries capacitor reliably.

[0572]FIG. 92A illustrates an exemplary down-conversion system 9252,according to an embodiment of the present invention, that may be used toindicate potential points in a signal path where DC offset voltages maybe injected. Down-conversion system 9252 comprises a UFD module 9246, afilter 9248, an amplifier 9202, and an optional IF down-converter 9250.UFD module 9246 comprises a UFT module 9254. UFD module 9246down-converts an input RF signal 9256, as described elsewhere herein,and outputs a down-converted signal 9258. Down-converted signal 9258 maybe a baseband signal, in which case IF down-converter 9250 is notrequired, or may be an intermediate frequency signal. Filter 9248receives and filters down-converted signal 9258, and outputs a filteredsignal 9260. Amplifier 9202 receives and amplifies filtered signal 9260,and outputs an amplified signal 9262. Optional IF down-converter 9250,when present, receives and further down-converts amplified signal 9262,and outputs an output signal 9264. Additional IF down-converter modulesmay be included as needed.

[0573] UFD module 9246, filter 9248, amplifier 9202, and optional IFdown-converter 9250 may each add a DC offset voltage to their respectiveoutputs signals. As described above, the DC offset voltage mayundesirably affect the value of the down-converted signal. It would bedesirable to provide a circuit that may be inserted for any of thecomponents shown in FIG. 92A (such as following such components), andany other applicable circuit components, to eliminate DC offset voltagesat that point.

[0574]FIG. 92B illustrates an exemplary auto-zero compensation circuit9200 for reducing or eliminating DC offset inserted by any of the abovedescribed circuit components, with amplifier 9202 of FIG. 92A shown asan example, according to an embodiment of the present invention. Thepresent invention is also applicable to reducing or eliminating DCoffsets inserted by other types of circuit components.

[0575] In the example circuit shown, auto-zero compensation circuit 9200is located following amplifier 9202 (see also FIG. 92A). In otherimplementations, auto-zero compensation circuit 9200 may follow anyapplicable circuit component in the down-converted signal path,including a UFD module. While amplifier 9202 is located in thedown-converted signal path, components of auto-zero compensation circuit9200 are located largely outside of the down-converted signal path.Auto-zero compensation circuit 9200 provides many of the same advantagesas having a capacitor located in series in the down-converted signalpath. Auto-zero compensation circuit 9200 comprises a resistor 9204, aswitch 9206, a capacitor 9208, a first summer 9210, a second summer9212, a first voltage reference 9214, and a second voltage reference9216.

[0576] Amplifier 9202 receives an input signal 9260, and outputs anamplified input signal 9262. Amplified input signal 9262 may comprise anunwanted DC offset voltage due to amplifier 9202. While an idealamplifier has zero input offset voltage (i.e., DC offset voltagereferred to the input) and no offset voltage drift, most actualamplifiers have offset voltages due to a mismatch of input transistorsand resistors on the monolithic circuit. This input offset voltage maydrift across temperature, and hence most amplifiers are specified withan input offset voltage temperature coefficient. An amplifier may sufferfrom further offset voltage from input bias currents. While an idealamplifier has zero current flowing into and out of its inputs, mostactual amplifiers have non-zero input bias currents flowing into and outof their inputs. These currents can create an input voltage thatresembles a DC offset voltage when they flow through resistors coupledto the amplifier inputs. Auto-zero compensation circuit 9200 removes DCoffset voltages and voltage drift created by these mechanisms.

[0577] A first terminal 9226 of resistor 9204 and a first terminal 9228of switch 9206 are coupled to amplified input signal 9262. A secondterminal 9230 of resistor 9204 and a second terminal 9232 of switch 9206are coupled to a first terminal 9234 of capacitor 9208 and a first inputterminal 9236 of second summer 9212. A third terminal 9244 of switch9206 is coupled to a receive mode signal 9242. A second terminal 9238 ofcapacitor 9208 is coupled to first voltage reference 9214. A secondinput terminal 9240 of second summer 9212 is coupled to second voltagereference 9216. First and second voltage references 9214 and 9216 may ormay not be equal to the same voltage value.

[0578] A receiver system may incorporate one or more auto-zerocompensation circuits 9200 in its down-converted signal path. When sucha receiver system enters a receive mode, i.e., it has entered a modewhere it is ready to down-convert received signals, a receive modesignal 9242 is activated. Receive mode signal 9242 causes switch 9206 toclose, and capacitor 9208 charges to the output voltage of amplifier9202. Hence, capacitor 9208 attains, or is charged with the value of theoutput of amplifier 9202, which comprises any DC offset voltage due toamplifier 9202. Capacitor 9208 may be a relatively large valuecapacitor, such as 1.0-0.1 μF, but the invention is not limited to thisrange.

[0579] After switch 9206 is closed for a length of time sufficient tocharge capacitor 9208 to the value of amplified output signal 9262, (theoutput of amplifier 9202), receive mode signal 9242 causes switch 9206to open. When switch 9206 is open, the path from amplifier 9202 to thefirst terminal 9234 of capacitor 9208 is through resistor 9204. Resistor9204 may be a relatively large value resistor, but the invention is notlimited to this example. In this configuration, capacitor 9208relatively slowly follows the voltage of the output of amplifier 9202.In this way, capacitor 9208 maintains the DC offset voltage value ofamplifier 9202, following any DC offset voltage drift due to changes inenvironmental temperature and the like.

[0580] Second summer 9212 adds the voltage stored in capacitor 9208 withthe value of second voltage reference 9216, and outputs adjusted DCoffset voltage 9224. Second voltage reference 9216 may be used to adjustor center the circuit output voltage, as described below. In alternateembodiments, second voltage reference 9216 and second summer 9212 arenot present, and the first terminal 9234 of capacitor 9208 is coupled tofirst summer 9210.

[0581] First summer 9210 subtracts the adjusted DC offset voltage 9224from amplified input signal 9262, and outputs DC offset adjusted outputsignal 9220 (which is received by the IF down-converter 9250 in theexample of FIG. 92A). DC Offset adjusted output signal 9220 issubstantially equal to input signal 9260 amplified by amplifier 9202,centered according to second voltage reference 9216 (if present), withthe DC offset due to amplifier 9202 substantially reduced or eliminated.

[0582]FIG. 110 depicts a flowchart 11000 that illustrates operationalsteps corresponding to FIG. 92B, for reducing DC offset in a signalpath, according to an embodiment of the present invention. The inventionis not limited to this operational description. Rather, it will beapparent to persons skilled in the relevant art(s) from the teachingsherein that other operational control flows are within the scope andspirit of the present invention. In the following discussion, the stepsin FIG. 110 will be described.

[0583] In step 11002, a DC offset voltage in an input signal is storedwhile in a signal non-receive mode.

[0584] In step 11004, the mode is changed to a signal receive mode.

[0585] In step 11006, the input signal is followed relatively slowly tomaintain the DC offset voltage and any DC offset voltage drift.

[0586] In step 11008, the maintained DC offset voltage is summed with acentering voltage to form an adjusted DC offset voltage signal. Acentering voltage such as second voltage reference 9216 may be used.

[0587] In step 11010, the adjusted DC offset voltage signal issubtracted from the input signal to form a DC offset adjusted outputsignal.

[0588] It should be understood that the above examples are provided forillustrative purposes only. The invention is not limited to thisembodiment. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. The invention is intended and adapted toinclude such alternate embodiments.

[0589] 7.2.1.5 Reducing DC Offset with Differential Configurations

[0590] DC offset voltages due to charge injection may also be reduced oreliminated through the use of differential UFD module configurations.Furthermore, circuit re-radiation may be reduced or eliminated throughthe use of differential UFD module configurations. Exemplarydifferential UFD module circuit embodiments are described below.However, it should be understood that these examples are provided forillustrative purposes only. The invention is not limited to theseembodiments. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. The invention is intended and adapted toinclude such alternate embodiments.

[0591] In an embodiment, two UFD modules are arranged in a differentialconfiguration, where a first UFD module receives an actual RF signal asan input, and a second UFD module receives circuit ground, or some othercircuit voltage, as an input. Furthermore, both UFD modules receive thesame control signal. As a result, the two UFD modules producesubstantially similar DC offset voltages due to charge injection. TheUFD module output signals may be subtracted from each other, and as aresult the DC offset voltage due to charge injection in the output ofthe first UFD module will be subtracted out.

[0592]FIG. 93 illustrates an exemplary differential DC offset voltagecancellation circuit 9300, according to an embodiment of the presentinvention. Differential DC offset voltage cancellation circuit 9300includes an optional LNA 9302, a first UFD module 9358, a second UFDmodule 9360, a control signal generator 9310, a dummy impedance 9312, asecond voltage reference 9314, and a summer 9322. In an embodiment,first UFD module 9358 comprises a first UFT module 9304, a first voltagereference 9306, and a first capacitor 9308, and second UFD module 9360comprises a second UFT module 9316, a second capacitor 9318, and a thirdvoltage reference 9320.

[0593] Optional LNA 9302 receives an input RF signal 9324 and outputs anamplified input RF signal 9326.

[0594] Amplified input RF signal 9326 is received by a first terminal9356 of first UFT module 9304. A second terminal 9338 of first UFTmodule 9304 is coupled to a first terminal 9340 of first capacitor 9308.First capacitor 9308 may be any type of applicable storage device. Athird terminal 9342 of first UFT module 9304 receives a control signal9328. Control signal 9328 is generated by control signal generator 9310.First UFT module 9304 down-converts amplified input RF signal 9326according to control signal 9328 in a manner as described elsewhereherein. First UFT module 9304 outputs actual output signal 9330 (it iscalled the “actual” output signal 9330 because it is derived from inputRF signal 9324), which is stored on first capacitor 9308. As describedabove, first UFT module 9304 may add unwanted DC offset voltage toactual output signal 9330, due to charge injection effects.

[0595] A first terminal 9344 of second UFT module 9316 receives dummyinput signal 9332 (it is called a “dummy” input signal 9332 because itis not a received signal, but is instead generated to address offsetissues) from a first terminal 9346 of dummy impedance 9312. A secondterminal 9348 of dummy impedance 9312 is coupled to second voltagereference 9314. Second voltage reference 9314 is a circuit voltage,preferably ground. Impedance 9312 approximates for second UFT module9316 the input impedance presented to the input of first UFT module 9304(that is, impedance 9312 is substantially equal to the input impedanceof first UFT module 9304). Impedance 9312 is implemented using any wellknown combination of circuit elements. A second terminal 9350 of secondUFT module 9316 is coupled to a first terminal 9352 of second capacitor9318. A third terminal 9354 of second UFT module 9316 receives controlsignal 9328. Second UFT module 9316 down-converts dummy input signal9332 according to control signal 9328 in a similar fashion as describedabove. Second UFT module 9316 outputs dummy output signal 9334, which isstored on second capacitor 9318. Dummy output signal 9334 comprisesunwanted DC offset voltage due to charge injection effects in second UFTmodule 9316, similar to that generated by first UFT module 9304. The DCoffset voltages due to charge injection on actual output signal 9330 anddummy output signal 9334 are substantially similar due to the similarUFT module configurations.

[0596] Summer 9322 subtracts dummy output signal 9334 from actual outputsignal 9330, and outputs output signal 9336. Output signal 9336 is adown-converted version of input RF signal 9324, with DC offset due tocharge injection in UFT module 9304 substantially reduced or eliminatedby subtracting out the DC offset similarly created in UFT module 9316.

[0597] Preferably, the noise entering on first terminals 9356 and 9344of UFT modules 9304 and 9316 is matched. If the frequency spectrum ofthe noise entering first UFT module 9304 on input RF signal 9324 isdifferent than the noise entering second UFT module 9316 from secondvoltage reference 9314, the difference may show up on output signal9336. One example of where the noise spectrums may be different is whenthere is a filter on input RF signal 9324 prior to first UFT module9304, which filters out some noise frequencies. This difference may besolved, for example, by placing a similar filter at the input of secondUFT module 9316.

[0598]FIG. 111 depicts a flowchart 11100 that illustrates operationalsteps, corresponding to the structure of FIG. 93, for down-converting aninput signal and canceling DC offset voltages, according to anembodiment of the present invention. The invention is not limited tothis operational description. Rather, it will be apparent to personsskilled in the relevant art(s) from the teachings herein that otheroperational control flows are within the scope and spirit of the presentinvention. In the following discussion, the steps in FIG. 111 will bedescribed.

[0599] In step 11102, an input signal is received.

[0600] In step 11104, the input signal is frequency down-converted witha first universal frequency down-conversion module to an actualdown-converted signal.

[0601] In step 11106, a dummy input signal is received. In anembodiment, a dummy impedance is matched with the input impedance of aninput of the first universal frequency down-conversion module. Thematched dummy impedance is coupled to an input of the second universalfrequency down-conversion module to form the dummy input signal.

[0602] In step 11108, the dummy signal is frequency down-converted witha second universal frequency down-conversion module to a dummydown-converted signal.

[0603] In step 11110, the dummy down-converted signal is subtracted fromthe actual down-converted signal to form an output signal. DC offsetvoltages due to said first and said second universal frequencydown-conversion modules are canceled by the subtraction of step 11110,as further described above.

[0604]FIG. 94A illustrates a second exemplary differential DC offsetvoltage cancellation circuit 9400, according to an embodiment of thepresent invention. Differential DC offset voltage cancellation circuit9400 is effective at reducing or eliminating DC offset voltages due tocharge injection and at reducing or eliminating circuit re-radiation.Differential DC offset voltage cancellation circuit 9400 comprises abuffer/inverter 9402, a first UFD module 9434, a second UFD module 9436,a control signal generator 9410, and a summer 9422. In an embodiment,first UFD module 9434 comprises a first UFT module 9404, a first voltagereference 9406, and a first capacitor 9408, and second UFD module 9436comprises a second UFT module 9416, a second capacitor 9418, and asecond voltage reference 9420.

[0605]FIG. 94B illustrates example waveforms related to differential DCoffset voltage cancellation circuit 9400 of FIG. 94A, according to anembodiment of the present invention.

[0606] Buffer/inverter 9402 receives an input RF signal 9424. FIG. 94Bshows an example waveform for input RF signal 9424. Buffer/inverter 9402outputs a non-inverted amplified input RF signal 9412 and an invertedamplified input RF signal 9414. Buffer/inverter 9402 may comprise anycircuit component or equivalent that receives a single-ended signal andoutputs a differential signal, such as a differential driver.Non-inverted amplified input RF signal 9412 and inverted amplified inputRF signal 9414 are substantially similar signals, but inverted images ofeach other. FIGS. 94C and 94D show example waveforms for non-invertedamplified input RF signal 9412 and inverted amplified input RF signal9414, respectively. In the example of FIGS. 94C and 94D, buffer/inverter9402 has a gain of 2, and therefore the amplitudes of non-invertedamplified input RF signal 9412 (FIG. 94C) and inverted amplified inputRF signal 9414 (FIG. 94D) are two times greater than that of input RFsignal 9424 (FIG. 94B).

[0607] First and second UFT modules 9404 and 9416 operate similarly tofirst and second UFT modules 9304 and 9316 of FIG. 93. First UFT module9404 receives non-inverted amplified input RF signal 9412. First UFTmodule 9404 operates to down-convert non-inverted amplified input RFsignal 9412 according to a control signal 9428, which is output bycontrol signal generator 9410. FIG. 94E shows an example waveform forcontrol signal 9428. First UFT module 9404 outputs a non-inverted outputsignal 9430. Non-inverted output signal 9430 comprises DC offset voltagedue to charge injection effects in first UFT module 9404, as describedabove. FIG. 94F shows an example waveform for non-inverted output signal9430. In this example, amplified input RF signal 9412 is down-convertedto non-inverted output signal 9430 at a value of 0.4 Volts, with a DCoffset voltage of 0.1 Volts added, resulting in a total of 0.5 Volts.

[0608] Second UFT module 9416 receives inverted amplified input RFsignal 9414. Second UFT module 9416 down-converts inverted amplifiedinput RF signal 9414 according to control signal 9428, and outputsinverted output signal 9432. Inverted output signal 9432 comprises DCoffset voltage due to charge injection in second UFT module 9416. FIG.94G shows an example waveform for inverted output signal 9432. Due atleast in part to the similarity in the layouts and circuitconfigurations of first and second UFT modules 9404 and 9416, theirresulting DC offset voltages due to charge injection will besubstantially similar, and of the same polarity. In the example of FIG.94G, inverted amplified input RF signal 9414 is down-converted toinverted output signal 9432 at a value of −0.4 Volts, with a DC offsetvoltage of 0.1 Volts added, resulting in a total of −0.3 Volts. As shownin FIGS. 94F and 94G, the polarities of non-inverted output signal 9430and inverted output signal 9432 are opposite. The DC offset voltagesadded respectively to these signals by first UFT module 9404 and secondUFT module 9416 are equal at 0.1 Volts.

[0609] Summer 9422 subtracts inverted output signal 9432 fromnon-inverted output signal 9430, and outputs an output signal 9426. FIG.94H shows an example waveform for output signal 9426. Becausenon-inverted output signal 9430 and inverted output signal 9432 comprisethe same down-converted signal, but of opposite polarities, whensubtracted by summer 9422, the respective down-converted signals willadd. Because the DC offset voltages in non-inverted output signal 9430and inverted output signal 9432 are of the same polarity and ofsubstantially the same amplitude, when they are subtracted by summer9422 the DC offset voltages will substantially cancel. As a result, anyDC offset voltage in output signal 9426 will be substantially reduced oreliminated. As shown in the example of FIGS. 94F-94H, the amplitudes ofnon-inverted output signal 9430 and inverted output signal 9432 combinein summer 9422 to equal 0.8 Volts, while the DC offset voltages of 0.1Volts cancel each other. Thus, the embodiment of FIG. 94A both enhancessignal amplitude and addresses DC offset issues.

[0610] Additionally, re-radiation may be substantially reduced oreliminated due to this configuration. Control signal noise produced infirst and second UFT modules 9404 and 9416 due to pulses on controlsignal 9428 may travel back through buffer/inverter 9402. If first andsecond UFT modules 9404 and 9416 are configured in a substantiallysimilar fashion and receive the same control signal, they will producesubstantially equivalent control signal noise. Because the controlsignal noise from second UFT module 9416 will be inverted bybuffer/inverter 9402 when passing back through buffer/inverter 9402, itwill cancel when combined with the non-inverted control signal noisefrom first UFT module 9404 passing back through buffer/inverter 9402.Furthermore, the noise matching concerns of the prior differentialcircuit embodiment of FIG. 93 are not present in this embodiment.

[0611]FIG. 112 depicts a flowchart 11200 that illustrates operationalsteps, corresponding to the structure of FIG. 94A, for down-convertingan input signal and canceling DC offset voltages, according to anembodiment of the present invention. The invention is not limited tothis operational description. Rather, it will be apparent to personsskilled in the relevant art(s) from the teachings herein that otheroperational control flows are within the scope and spirit of the presentinvention. In the following discussion, the steps in FIG. 112 will bedescribed.

[0612] In step 11202, an input signal is received.

[0613] In step 11204, the received input signal is amplified to anon-inverted output signal and an inverted output signal.

[0614] In step 11206, the non-inverted output signal is down-convertedwith a first universal frequency down-conversion module to anon-inverted down-converted signal.

[0615] In step 11208, the inverted output signal is down-converted witha second universal frequency down-conversion module to an inverteddown-converted signal.

[0616] In step 11210, the inverted down-converted signal is subtractedfrom the non-inverted down-converted signal to form an output signal. DCoffset voltages in the non-inverted down-converted signal and theinverted down-converted signal produced by the first and seconduniversal frequency down-conversion modules, respectively, are canceled.

[0617] In step 11212, the first universal frequency down-conversionmodule and the second universal frequency down-conversion module areconfigured to generate substantially equal DC offset voltages given thesame input signal.

[0618] It should be understood that the above examples are provided forillustrative purposes only. The invention is not limited to theseembodiments. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. The invention is intended and adapted toinclude such alternate embodiments.

[0619] 7.2.1.6 Reducing DC Offset with Differential Outputs

[0620] Unwanted DC offset voltages may be reduced or canceled throughthe use of differential receiver circuit outputs. FIG. 95 illustrates anexemplary differential receiver circuit 9500, according to an embodimentof the present invention. Differential receiver circuit 9500 comprises afirst impedance match 9502, a second impedance match 9504, a tankcircuit 9506, a differential UFD module 9508, a control signal generator9510, and a resistor 9512.

[0621] First and second impedance match 9502 and 9504 are optional, thenecessity of which being determined on an application-by-applicationbasis. In a preferred embodiment, first impedance match 9502 is a firstinductor 9514. In a preferred embodiment, second impedance match 9504 isa second inductor 9516. However, other impedance match circuits may beused.

[0622] Tank circuit 9506 is optional, the necessity of which beingdetermined on an application-by-application basis. In a preferredembodiment, tank circuit 9506 comprises a first capacitor 9518 and athird inductor 9520, although other circuits may be used.

[0623] In a preferred embodiment, differential UFD module 9508 comprisesa first UFT module 9522, a second UFT module 9524, and a storage module9534. In a preferred embodiment, storage module 9534 comprises a secondcapacitor 9526.

[0624] A positive or “plus” signal input of a differential RF inputsignal 9528 is input through first impedance match 9502 to a firstterminal 9536 of tank circuit 9506. A negative or “minus” signal inputof differential RF input signal 9528 is input through second impedancematch 9504 to a second terminal 9538 of tank circuit 9506.

[0625] First UFT module 9522 is coupled to first terminal 9536 oftankcircuit 9506, and receives the “plus” signal input of differential RFinput signal 9528. Second UFT module 9524 is coupled to second terminal9538 of tank circuit 9506, and receives the “minus” signal input ofdifferential RF input signal 9528.

[0626] First and second UFT modules 9522 and 9524 down-convertdifferential RF input signal 9528 according to a control signal 9532,which is output by control signal generator 9510, in a manner asdescribed elsewhere herein. The outputs of first and second UFT modules9522 and 9524 are stored in storage module 9534, and output asdifferential output signal 9530.

[0627] First UFT module 9522 outputs a “plus” output of differentialoutput signal 9530. Second UFT module 9524 outputs a “minus” output ofdifferential output signal 9530. Differential output signal 9530 isequal to the difference voltage between these “plus” and “minus”outputs.

[0628] A first terminal 9540 of storage module 9534 is coupled to the“plus” output of differential output signal 9530. A second terminal 9542of storage module 9534 is coupled to the “minus” output of differentialoutput signal 9530.

[0629] Resistor 9512 is optional, the necessity of which beingdetermined on an application-by-application basis. Resistor 9512, whenpresent, operates as a load resistance, the value of which may bedetermined on an application-by-application basis. A first terminal 9544of resistor 9512 is coupled to the “plus” output of differential outputsignal 9530. A second terminal 9546 of resistor 9512 is coupled to the“minus” output of differential output signal 9530.

[0630] Due to their similar layout and circuit configuration, and due tocontrol signal 9532, first UFT module 9522 and second UFT module 9524each generate substantially equal DC offset voltages due to chargeinjection effects. The DC offset voltage generated by first UFT module9522 is applied to first terminal 9540 of storage module 9534. The DCoffset voltage generated by second UFT module 9524 is applied to secondterminal 9542 of storage module 9534. Because differential output signal9532 is measured across storage module 9534, the DC offset voltages dueto first and second UFT module 9522 and 9524 substantially cancel eachother out.

[0631]FIG. 113 depicts a flowchart 11300 that illustrates operationalsteps, corresponding to the structure of FIG. 95, for differentiallydown-converting an input signal, according to an embodiment of thepresent invention. The invention is not limited to this operationaldescription. Rather, it will be apparent to persons skilled in therelevant art(s) from the teachings herein that other operational controlflows are within the scope and spirit of the present invention. In thefollowing discussion, the steps in FIG. 113 will be described.

[0632] In step 11302, an input signal is differentially received. Forexample, a positive node input signal and a negative node input signalare received.

[0633] In step 11304, the differentially received input signal isdown-converted with a differential universal frequency down-conversionmodule to a differential down-converted signal. The differentialdown-converted signal comprises a positive node down-converted signaland a negative node down-converted signal. In an embodiment, thedifferential universal frequency down-conversion module comprises apositive node switch (UFT) module and a negative node switch (UFT)module. The positive node switch module and the negative node switchmodule are configured to generate substantially equal DC offset voltagesin the positive node down-converted signal and the negative nodedown-converted signal, respectively, as described above.

[0634] In step 11306, the differential down-converted signal is measuredbetween the positive node down-converted signal and the negative nodedown-converted signal. The DC offset voltages in the positive nodedown-converted signal and the negative node down-converted signalsubstantially cancel, as described above.

[0635] Further differential circuit configurations for canceling DCoffset voltages will be apparent to persons skilled in the relevantart(s) from the teaching herein. Exemplary differential receiver circuitoutput embodiments are described above. However, it should be understoodthat these examples are provided for illustrative purposes only. Theinvention is not limited to these embodiments. Alternate embodiments(including equivalents, extensions, variations, deviations, etc., of theembodiments described herein) will be apparent to persons skilled in therelevant art(s) based on the teachings contained herein. The inventionis intended and adapted to include such alternate embodiments.

[0636] 7.2.2 Re-Radiation

[0637] Re-radiation, as described above, is an undesirable phenomenonwhere a signal comprising one or more frequency components generated byreceiving circuitry is transmitted by an antenna. FIG. 44A illustratesan antenna 4406 that transmits circuit re-radiation signals 4414 and4420. Receiving circuitry, for example shown as a receiver 4402 and alocal oscillator 4404, related to antenna 4406 may produce thetransmitted frequency components. For example, the frequency componentsmay be generated in part by local oscillator 4404. These generatedfrequency components may travel along re-radiation path 4418, where theyare transmitted by antenna 4406 as re-radiation signals 4414 and 4420.When transmitted, these frequency components may undesirably interferewith one or more nearby receivers, such as nearby receiver 4408. Anantenna 4420 may receive re-radiation signal 4420, which isdown-converted by nearby receiver 4408. One or more of the frequencycomponents of received re-radiation signal 4420 may fall within afrequency range of interest of nearby receiver 4408, interfering withthe quality of the signals intended to be down-converted by nearbyreceiver 4408.

[0638] As described above, re-radiation may be undesirably received backby the same antenna that transmitted the re-radiation. As shown in FIG.44A, receiver 4402 may transmit re-radiation signal 4414, which issubsequently reflected by an object 4412 as reflected re-radiation 4416,which is then received by antenna 4406. This is referred to asre-radiation recapture. If frequency components are received back by thesame antenna that transmitted them, they may be down-converted, mayfurther impair signals that are down-converted, and/or may causeundesirable DC offset voltages that may impair the down-convertedsignals. FIG. 44B shows an example local oscillator signal 4422, of afrequency of f. If a signal such as local oscillator signal 4422 isre-radiated, and subsequently received by the circuit that transmittedit, it may combine with itself to create an undesired DC offset voltage.FIG. 44C shows the Fourier transform of local oscillator signal 4422,with spectral components 4424 and 4426 at frequencies +f and −f. FIG.44D shows a result of the convolution of local oscillator signal 4422with itself, producing a DC spectral component 4432 representing anundesired DC offset voltage. FIG. 44D shows resulting spectralcomponents 4428 and 4430 at frequencies +2f and −2f, and DC spectralcomponent 4432 at a frequency of zero. DC spectral component 4432 maycause the same problems as DC offset voltages created by othermechanisms, such as those described elsewhere herein.

[0639] For at least these reasons, it is desirable to reduce oreliminate circuit re-radiation. Exemplary embodiments are provided belowfor reducing or eliminating circuit re-radiation. The embodimentsprovided below are not limited to this use, but may have additionalapplications. For example, these embodiments may be applicable toreducing or eliminating unwanted DC offset voltages.

[0640] 7.2.2.1 Reducing Re-Radiation by Adjusting Control SignalAttributes

[0641] In the present invention, a local oscillator may be used togenerate a control signal used to down-convert received RF signals. Thecontrol signal may comprise frequency components related to the localoscillator frequency and its harmonics. As described above, one or morefrequency components of the local oscillator signal may leak from anearby antenna as circuit re-radiation. As a result, attributes ofcircuit re-radiation are directly related to attributes of controlsignal frequency components. Hence, re-radiation potentially may bereduced or eliminated by adjusting one or more attributes of the controlsignal frequency components. Control signal attributes that may beadjusted at least include control pulses width, control pulse amplitude,and/or control pulse phase.

[0642]FIG. 96 shows an exemplary input RF signal 9602. A π-pulse lengthcontrol signal 9604 is also shown that may be applied to a UFD module todown-convert input RF signal 9602. As shown, π-pulse length controlsignal 9604 comprises pulses that are of a length of π radians. In areceiver embodiment implementing a UFD module, a control signal such asπ-pulse length control signal 9604 may be re-radiated from the receiver.In the time domain, the re-radiation may appear as noise pulses that areshaped similarly to pulses of the control signal. In certain situations,one or more of the frequencies of the re-radiated signal may undesirablyfall within the output frequency bands of interest of the systemimplementing the receiver circuit. For instance, when down-converting asignal directly to baseband, a control signal frequency may besubstantially equal to the frequency of the received RF carrier signal.If this control signal frequency is re-radiated, and then subsequentlyreceived and down-converted, it may result in one or more down-convertedsignal frequencies near or equal to DC, or at baseband, in a similarfashion to that described in FIGS. 44B-D above. It would be beneficialif the re-radiated signal components within the frequency bands ofinterest could be eliminated or moved.

[0643] In an exemplary embodiment for changing the frequency content ofthe re-radiated signal, the pulse width of the control pulses of thecontrol signal may be lengthened. As shown in FIG. 96, a 3π-pulse lengthcontrol signal 9606 has control pulses of a length of 3π. Because thecontrol pulse width of 3π-pulse length control signal 9606 is wider thanthat of π-pulse length control signal 9604, 3π-pulse length controlsignal 9606 is made up of lower frequency components. It is well knownthat signals comprising substantially square or rectangular pulsesinclude a plurality of signals of various frequencies that add togetherto form the pulse shapes. As pulses become wider, the frequencies of thesignals required to form them tend to become lower. Because 3π-pulselength control signal 9606 has wider pulses, and therefore containslower frequency components, a re-radiated signal due to 3π-pulse lengthcontrol signal 9606 will have lower frequency components. Even if thelower frequency components are re-radiated, and then received anddown-converted, the down-converted components should be out-of-band. Inan embodiment, a 3π-pulse length control signal 9606 configurationre-radiated at a 19 dB lower level than that of a π-pulse length controlsignal 9604.

[0644] Frequency components of potential re-radiation can be loweredmore by further widening the control pulses. For example, FIG. 96 showsa 5π-pulse length control signal 9608 with control signal pulses of awidth of 5π. 5π-pulse length control signal 9608 includes pulses widerthan those of 3π-pulse length control signal 9606. Because of this, asdescribed above, 5π-pulse length control signal 9608 is made up of lowerfrequency signal components relative to 3π-pulse length control signal9606. Hence, relative to 3π-pulse length control signal 9606, circuitre-radiation related to 5π-pulse length control signal 9608 is of lowerfrequency.

[0645] A pulse width can be widened even more as would be understood bypersons skilled in the relevant arts from the teachings herein. To whatdegree the pulse width may be widened will be determined on anapplication by application basis. The pulse width may be varied by wholeincrements of π, or any fraction thereof. It should be understood thatthe above pulse width examples are provided for illustrative purposesonly. The invention is not limited to these embodiments. Alternateembodiments (including equivalents, extensions, variations, deviations,etc., of the embodiments described herein) will be apparent to personsskilled in the relevant art(s) based on the teachings contained herein.The invention is intended and adapted to include such alternateembodiments.

[0646]FIG. 114 depicts a flowchart 11400 that illustrates operationalsteps for down-converting an input signal with a variety of controlsignal pulse widths, according to an embodiment of the presentinvention. The invention is not limited to this operational description.Rather, it will be apparent to persons skilled in the relevant art(s)from the teachings herein that other operational control flows arewithin the scope and spirit of the present invention. In the followingdiscussion, the steps in FIG. 114 will be described.

[0647] In step 11402, an input signal is frequency down-converted with auniversal frequency down-conversion module to a down-converted signal.The input signal is down-converted according to a control signalcomprising a train of pulses having pulse widths.

[0648] In step 11404, a signal related to the control signal isre-radiated.

[0649] In step 11406, the pulse widths are increased to decrease afrequency of the re-radiated signal. In an embodiment, the pulse widthsmay be selected according to the equation: pulse width=180+360·n degreesof a frequency of said input signal, wherein n is any integer≧zero. As nis increased, a frequency of the re-radiated signal is decreased.

[0650] 7.2.2.1.1 I/Q Modulation Receiver Control Signal Considerationsand Embodiments

[0651] Design considerations exist for I/Q modulation receiver circuitsin regard to control signals. The embodiments provided above forchanging control signal pulse widths are applicable to I/Q modulationreceiver circuits. However, when modifying control signal pulse widthsin regards to I/Q modulation receiver circuits to overcome problems withre-radiation as described above, or other problems, certain designconstraints may need to be considered. For instance, in someembodiments, such as described below, pulses of the I-phase controlsignal and pulses of the Q-phase control signal may not overlap, andmust be configured such that they do not overlap to fulfill thisrequirement. In alternate embodiments, such as described below, an I/Qmodulation receiver circuit may be configured such that I-phase andQ-phase control signals may overlap. Exemplary embodiments are providedbelow for overcoming at least some design constraints related to controlsignal pulses for I/Q modulation receiver circuits, according to thepresent invention.

[0652] It should be understood that the following I/Q modulationreceiver examples are provided for illustrative purposes only. Theinvention is not limited to these embodiments. Alternate embodiments(including equivalents, extensions, variations, deviations, etc., of theembodiments described herein) will be apparent to persons skilled in therelevant art(s) based on the teachings contained herein. The inventionis intended and adapted to include such alternate embodiments.

[0653] 7.2.2.1. 1.1 Non-overlapping I/Q Control Signal PulsesEmbodiments

[0654]FIG. 97 illustrates an exemplary I/Q modulation receiver circuit9700, according to an embodiment of the present invention. I/Qmodulation receiver circuit 9700 comprises a first UFD module 9702, asecond UFD module 9704, a control signal generator 9706, and a phaseshifter 9708. I/Q modulation circuit 9700 may use a variety of controlsignal configurations to down-convert I/Q modulated signals.

[0655] An input RF I/Q signal 9722 is received by first UFD module 9702.First UFD module 9702 down-converts the I-phase signal portion of inputRF I/Q signal 9722 according to a control signal 9728, which is outputby control signal generator 9706. First UFD module 9702 outputs an Ioutput signal 9724.

[0656] In an embodiment, first UFD module 9702 comprises a first UFTmodule 9710, a first storage module 9712, and a first voltage reference9714.

[0657] Control signal 9728 is received by phase shifter 9708. In an I/Qmodulation embodiment, phase shifter 9708 preferably shifts the phase ofcontrol signal 9728 by 90 degrees, although other phase shifts arepossible. Phase shifter 9708 outputs phase-shifted control signal 9730.

[0658] Input RF I/Q signal 9722 is received by second UFD module 9704.Second UFD module 9704 down-converts the Q-phase signal portion of inputRF I/Q signal 9722 according to phase-shifted control signal 9730.Second UFD module 9704 outputs a Q output signal 9726.

[0659] In an embodiment, second UFD module 9704 comprises a second UFTmodule 9716, a second storage module 9718, and a second voltagereference 9720. First and second voltage references 9714 and 9720 may ormay not be equal to the same voltage value.

[0660] FIGS. 98A-981 shows an exemplary input RF I/Q signal 9722, andseveral exemplary control signal waveforms, which may be used todown-convert input RF I/Q signal 9722.

[0661] For example, an I-control signal 9802 is shown in FIG. 98B.I-control signal 9802 may be used to down-convert an I-phase signalportion of input RF I/Q signal 9722. A corresponding Q-control signal9804 is shown in FIG. 98C. Q-control signal 9804 is output by phaseshifter 9708. Q-control signal 9804 is shifted by 90 degrees fromI-control signal 9802. Q-control signal 9804 may be used to down-converta Q-phase signal portion of input RF I/Q signal 9722.

[0662] As illustrated in FIGS. 98B and 98C, pulses of I-control signal9802 overlap the corresponding phase-shifted pulses of Q-control signal9804. In some embodiments where first and second UFT modules 9710 and9716 comprises switches, overlapping pulses of I-control signal 9802 andQ-control signal 9804 will cause the switches in first and second UFTmodules 9710 and 9716 to be simultaneously closed during a period pulseof overlap. Due to the overlap, first and second UFD modules 9702 and9704 may not be able to properly down-convert the I- and Q-phasecomponents of input RF I/Q signal 9722. This is because during theperiod that switches inside the first and second UFD modules 9702 and9704 are both closed, the switches will be attempting to transfer energyfrom input RF I/Q signal 9722 simultaneously. This may lead tonon-negligible distortion of input RF I/Q signal 9722 in someembodiments. Hence, less than desirable input signal down-conversionaccuracy may result.

[0663] In an another example, FIGS. 98D and 98E show a 3π I-controlsignal 9806 and a 3π Q-control signal 9808. Pulses of 3π I-controlsignal 9806 and 3π Q-control signal 9808 overlap. Using these controlsignals, in some embodiments first and second UFD modules 9702 and 9704may not be able to properly down-convert the I- and Q-phase componentsof input RF I/Q signal 9722.

[0664] The overlap problem may be overcome by creating control signalswith non-overlapping pulses. For example, FIGS. 98F and 98G show anon-overlapping I-control signal 9810 and a non-overlapping Q-controlsignal 9812. Pulses on non-overlapping I-control signal 9810 areseparated by 720 degrees, and may be used to down-convert the I-phasesignal component of input RF I/Q signal 9722. Pulses on non-overlappingQ-control signal 9812 are phased-shifted from pulses on non-overlappingI-control signal 9810 by 270 degrees, are separated from each other by720 degrees, and may be used to down-convert the Q-phase signalcomponent of input RF I/Q signal 9722.

[0665] In a further example, FIGS. 98H and 98I show a non-overlappingI-control signal 9814 and anon-overlapping Q-control signal 9816. Pulseson non-overlapping I-control signal 9814 are separated by 540 degrees,and may be used to down-convert the I-phase signal component of input RFI/Q signal 9722. (Note that when pulses on non-overlapping I-controlsignal 9814 are separated by 180 degrees, 540 degrees, 900 degrees,etc., the information down-converted on consecutive pulses may beinverted relative to one another, and hence an inverter may be requiredto correct for this.) Pulses on non-overlapping Q-control signal 9816are phased-shifted from pulses on non-overlapping I-control signal 9814by 270 degrees, are separated from each other by 540 degrees, and may beused to down-convert the Q-phase signal component of input RF I/Q signal9722. (Note that when pulses on non-overlapping Q-control signal 9816are separated by 180 degrees, 540 degrees, 900 degrees, etc., theinformation down-converted on consecutive pulses may be invertedrelative to one another, and hence an inverter may be required tocorrect for this.)

[0666] Further control signal waveform configurations exist forimplementing non-overlapping pulses, according to embodiments of thepresent invention, as would be recognized by persons skilled in therelevant art(s) from the teachings herein. I- and Q-control signalpulses may be widened, or made more narrow. I- and Q- control signalpulses may be made to occur further apart or closer together. AQ-control signal may be phase-shifted from a corresponding I-controlsignal by 90 degrees, 270 degrees, 450 degrees, 630 degrees, and so on,such that the I-control signal is matched with the I-phase input RFsignal component, and the Q-control signal is matched with the Q-phaseinput RF signal component. Pulses on an I-phase control signal may beshifted from each other by any multiple of 180 (with one or moreinverters possibly required, as described above) or 360 degrees. I- andQ-control signals may be formed to these requirements for use in I/Qmodulation receiver circuit 9700 of FIG. 97, as long as their pulses donot overlap.

[0667]FIG. 115 depicts a flowchart 11500 that illustrates operationalsteps corresponding to the structure of FIG. 97, for down-converting anRF I/Q modulated input signal, according to an embodiment of the presentinvention. The invention is not limited to this operational description.Rather, it will be apparent to persons skilled in the relevant art(s)from the teachings herein that other operational control flows arewithin the scope and spirit of the present invention. In the followingdiscussion, the steps in FIG. 115 will be described.

[0668] In step 11502, an input RF I/Q modulated signal is frequencydown-converted with a first universal frequency down-conversion moduleaccording to a control signal. The input signal is down-converted to anin-phase information signal. The control signal comprises a train ofpulses. In an embodiment, the train of pulses are generated to haveapertures approximately equal to 180+360·n degrees of a frequency ofsaid input RF I/Q modulated signal, wherein n is any integer greaterthan or equal to 0.

[0669] In step 11504, the control signal is phase-shifted. Inembodiments, the control signal is phase-shifted by 90 degrees of afrequency of said input RF I/Q modulated signal. In alternativeembodiments, the control signal may be shifted by 90+m·180 degrees,wherein m is any integer greater than or equal to 1. In embodiments, thecontrol signal may be phase shifted such that pulses on the controlsignal do not overlap pulses on the phase-shifted control signal.

[0670] In step 11506, the input RF I/Q modulated signal is frequencydown-converted with a second universal frequency down-conversion moduleaccording to the phase-shifted control signal. The input signal isdown-converted to a quadrature-phase information signal.

[0671] 7.2.2.1.1.2 Buffered I/Q Modulation Receiver Embodiment

[0672] Exemplary embodiments are provide below for I/Q modulationreceiver circuits where control signal pulses may overlap. Suchembodiments may provide advantages where it is desirable to modifycontrol signal pulse attributes as described above to solve problemswith circuit re-radiation, and other problems. Additional and alternateembodiments will be recognized by persons skilled in the relevant art(s)from the teachings herein, and are within the scope of the presentinvention.

[0673]FIG. 99 illustrates an exemplary buffered I/Q modulation receivercircuit 9900, according to an embodiment of the present invention.Buffered I/Q modulation receiver circuit 9900 allows for overlapping I-and Q-control signal pulses such as I-control signal 9802 and Q-controlsignal 9804 of FIG. 98.

[0674] Buffered I/Q modulation receiver circuit 9900 comprises anoptional splitter 9902, a first low noise amplifier (LNA) 9904, a secondLNA 9908, a control signal generator 9910, a first UFD module 9912, asecond UFD module 9914, and a phase shifter 9916. Buffered I/Qmodulation receiver circuit 9900 is configured substantially similar to,and operates in a similar fashion to I/Q modulation receiver circuit9700 of FIG. 97, with the addition of optional splitter 9902, first LNA9904, and second LNA 9908.

[0675] Optional splitter 9902 optionally splits an input RF I/Q signal9930, and outputs a first split input RF I/Q signal 9944 to first LNA9904, and a second split input RF I/Q signal 9946 to second LNA 9908.

[0676] First LNA 9904 buffers and optionally amplifies first split inputRF I/Q signal 9944, and outputs a first buffered input RF I/Q signal9936.

[0677] Second LNA 9908 buffers and optionally amplifies second splitinput RF I/Q signal 9946, and outputs a second buffered input RF I/Qsignal 9938.

[0678] First UFD module 9912 receives first buffered input RF I/Q signal9936. First UFD module 9912 down-converts first buffered input RF I/Qsignal 9936 according to a control signal 9940, which is output bycontrol signal generator 9910. First UFD module 9912 outputs I outputsignal 9932. In an embodiment, first UFD module 9912 comprises a firstUFT module 9918, a first storage module 9920, and a first voltagereference 9922.

[0679] Phase shifter 9916 receives control signal 9940, and outputs aphase-shifted control signal 9942. Phase-shifted control signal 9942 ispreferably shifted by 90 degrees from control signal 9940, but may alsobe shifted by 270 degrees, 450 degrees, 630 degrees, and so on.

[0680] Second UFD module 9914 receives second buffered input RF I/Qsignal 9938. Second UFD module 9914 down-converts second buffered inputRF I/Q signal 9938 according to phase-shifted control signal 9942.Second UFD module 9914 outputs Q output signal 9934. In an embodiment,second UFD module 9914 comprises a second UFT module 9924, a secondstorage module 9926, and a second voltage reference 9928.

[0681] As described elsewhere herein, when first UFT module 9918transfers energy from first buffered input RF I/Q signal 9936, firstbuffered input RF I/Q signal 9936 will be distorted to some degree.Likewise, when second UFT module 9924 transfers energy from secondbuffered input RF I/Q signal 9938, second buffered input RF I/Q signal9938 will be distorted to some degree. First and second LNA 9904 and9908 buffer the input RF I/Q signals entering first and second UFDmodules 9912 and 9914 from input RF I/Q signal 9930. Hence, input RF I/Qsignal 9930 will not be substantially distorted by energy transferoccurring in either of first and second UFD module 9912 and 9914.Because of this, the I- and Q-control signals used to cause first andsecond UFD modules 9912 and 9914 to down-convert their respective inputRF I/Q signals may have overlapping I- and Q-pulses. Hence, for example,control signal 9940 may appear as I-control signal 9802 of FIG. 98, andphase-shifted control signal 9942 may appear as Q-control signal 9804,phase-shifted by 90 degrees from, and having pulses overlapping with,control signal 9940. It is noted that the invention is not limited tothe example of FIG. 99. Other components to buffer or isolate first andsecond UFT modules 9918,9924 from each other could alternatively beused.

[0682]FIG. 116 depicts a flowchart 11600 that illustrates operationalsteps corresponding to the structure of FIG. 99, for down-converting anRF I/Q modulated input signal, according to an embodiment of the presentinvention. The invention is not limited to this operational description.Rather, it will be apparent to persons skilled in the relevant art(s)from the teachings herein that other operational control flows arewithin the scope and spirit of the present invention. In the followingdiscussion, the steps in FIG. 116 will be described.

[0683] In step 11602, an input RF I/Q modulated signal is buffered witha first low noise amplifier and a second low noise amplifier. In analternative embodiment, instead of or in addition to buffering the inputRF I/Q modulated signal as just described, the input RF I/Q modulatedsignal may be split into a first split RF I/Q modulated signal and asecond split RF I/Q modulated signal.

[0684] In step 11604, the first buffered (and/or first split) RF I/Qmodulated signal is frequency down-converted with a first universalfrequency down-conversion module according to a control signal. Theinput signal is down-converted to an in-phase information signal. Thecontrol signal comprises a train of pulses. In an embodiment, the trainof pulses are generated to have apertures approximately equal to180+360·n degrees of a frequency of said input RF I/Q modulated signal,wherein n is any integer greater than or equal to 0.

[0685] In step 11606, the control signal is phase-shifted. Inembodiments, the control signal is phase-shifted by 90+m·180 degrees ofa frequency of said input RF I/Q modulated signal, wherein m is anyinteger greater than or equal to 0. In embodiments, the control signalmay be phase shifted such that pulses on the control signal overlappulses on the phase-shifted control signal.

[0686] In step 11608, the second buffered (and/or second split) RF I/Qmodulated signal is frequency down-converted with a second universalfrequency down-conversion module according to the phase-shifted controlsignal. The input signal is down-converted to a quadrature-phaseinformation signal.

[0687] 7.2.2.2 Reducing Re-Radiation with Placebo Down-ConversionModules

[0688]FIG. 100 illustrates an exemplary receiver 10000 with placebocircuit 10004, according to an embodiment of the present invention.Receiver 10000 with placebo circuit 10004 reduces or frequency shiftspotentially re-radiated control signal components such that theirpotentially adverse impact on a down-converted signal is reduced. Thepotentially re-radiated control signal components may be shifted out ofthe frequency bands of interest, such that they will have a reducedadverse impact on the down-converted signal.

[0689] Control signal frequency components may be adjusted or shiftedthrough the use of one or more UFT modules, called “placebo” UFTmodules, and one or more corresponding “placebo” control signals. In aplacebo embodiment, an “actual” UFT module receives and down-converts areceived RF input signal with an “actual” control signal as describedelsewhere herein. Furthermore, a placebo UFT module receives a placebocontrol signal, and may also down-convert the received RF input signal,to output a down-converted signal. The actual control signal and one ormore placebo control signals may cause circuit re-radiation. Thisresulting circuit re-radiation will be related to a combination of theactual control signal waveform and the one or more placebo controlsignal waveforms. Hence, attributes of the resulting circuitre-radiation may be manipulated by using various placebo control signalwaveforms, to cause overall circuit re-radiation to be less harmful tocircuit performance. Characteristics of a particular placebo controlsignal waveform may be determined on an application-by-applicationbasis. The term “placebo” is used because the signal down-converted bythe placebo circuitry is not necessarily used by subsequent signalprocessing hardware and software, but may actually remain unutilized.The signal down-converted by the “actual” circuitry is used bysubsequent signal processing.

[0690] Receiver 10000 with placebo circuit 10004 comprises an actual UFDmodule 10002, a placebo UFD module 10004, a control signal generator10006, and a phase shifter 10008.

[0691] Actual UFD module 10002 receives an input RF signal 10022. ActualUFD module 10002 down-converts actual input RF signal 10022 according toa control signal 10028, which is output by control signal generator10006, in a manner as described elsewhere herein. Actual UFD module10002 outputs an actual output signal 10024. In an embodiment, actualUFD module 10002 comprises an actual UFT module 10010, an actual storagemodule 10012, and an actual voltage reference 10014.

[0692] Phase shifter 10008 receives control signal 10028, and outputs aphase-shifted placebo control signal 10030. Phase-shifted placebocontrol signal 10030 is preferably shifted such that pulses onphase-shifted placebo control signal 10030 do not overlap with pulses oncontrol signal 10028. In other embodiments, pulses on phase-shiftedplacebo control signal 10030 may overlap pulses on control signal 10028,as would be understood by persons skilled in the relevant art(s) fromthe teachings herein.

[0693] Placebo UFD module 10004 receives input RF signal 10022. PlaceboUFD module 10004 down-converts input RF signal 10022 according tophase-shifted placebo control signal 10030. Placebo UFD module 10004outputs placebo output signal 10026. In an embodiment, placebo UFDmodule 10004 comprises a placebo UFT module 10016, a placebo storagemodule 10018, and a placebo voltage reference 10020.

[0694]FIG. 101 shows an exemplary control signal waveform 10102, and acorresponding exemplary placebo control signal waveform 10104 that is adelayed (or phase-shifted) version of control signal waveform 10102.Control signal 10028 may comprise a control signal waveform such ascontrol signal waveform 10102. Phase-shifted placebo control signal10030 may comprise a corresponding control signal waveform such asplacebo control signal waveform 10104.

[0695] In a receiver circuit embodiment that does not include a placeboUFD module 10004, potential circuit re-radiation (and the frequencyspectrum of such re-radiation) will be related to the control signalwaveform being used, such as control signal waveform 10102. In areceiver circuit embodiment that includes a placebo UFD module 10004,the potential circuit re-radiation (and the frequency spectrum of suchre-radiation) will be related to the control signal being used, such ascontrol signal waveform 10102, and the placebo control signal waveformbeing used, such as placebo control signal waveform 10104.

[0696]FIG. 101 shows a combined signal waveform 10106 that represents acombination of control signal waveform 10102 and placebo control signalwaveform 10104. The potential circuit re-radiation (and the frequencyspectrum thereof) due to control signal waveform 10102 and placebocontrol signal waveform 10104 will be related to combined signalwaveform 10106 (and the frequency spectrum thereof). As would beapparent to persons skilled in the relevant art(s), combined signalwaveform 10106 has a different frequency spectrum than control signalwaveform 10102. By utilizing at least one placebo control signal inaddition to an actual control signal, a potentially re-radiatedfrequency spectrum can be adjusted to move potentially harmful circuitnoise and potentially resulting DC offset and/or re-radiated componentsto a non-critical frequency band in output signal 10024.

[0697] Furthermore, as shown in FIG. 101, placebo control signalwaveform 10104 may be phase-shifted by lesser or greater amounts fromcontrol signal waveform 10102. Arrows 10108, 10110, 10112, and 10114indicate possible variations in the phase of placebo control signalwaveform 10104, with the corresponding variations in combined signalwaveform 10106 indicated by arrows 10116, 10118, 10120, and 10122. Bychanging the phase of placebo control signal waveform 10104 in relationto control signal waveform 10102, the frequency spectrum of potentialre-radiation may be adjusted. Furthermore, changes in the amplitudeand/or width of pulses on placebo control signal waveform 10104 may alsobe used to adjust the frequency spectrum and amplitude of potentialre-radiation.

[0698] In embodiments, placebo output signal 10026 is not used indown-stream information signal processing. In alternative embodiments,placebo output signal 10026 may be used in down-stream informationsignal processing.

[0699]FIG. 117 depicts a flowchart 11700 that illustrates operationalsteps corresponding to the structure of FIG. 100 and waveforms of FIG.101, for down-converting an input signal and altering circuitre-radiation, according to an embodiment of the present invention. Theinvention is not limited to this operational description. Rather, itwill be apparent to persons skilled in the relevant art(s) from theteachings herein that other operational control flows are within thescope and spirit of the present invention. In the following discussion,the steps in FIG. 117 will be described.

[0700] In step 11702, an input signal is frequency down-converted with afirst universal frequency down-conversion module to a firstdown-converted signal, wherein the input signal is down-convertedaccording to a control signal, wherein the control signal comprises atrain of pulses, wherein pulses of the control signal occur every360+360·n degrees of a frequency of the input signal, wherein n is equalto any integer greater or equal 0.

[0701] In step 11704, the control signal is phase-shifted, wherein thecontrol signal is phase shifted in a range between 0 degrees and360+360·n degrees of a frequency of the input signal (pulses of controlsignal and phase-shifted control signal may overlap). In alternativeembodiments, the pulses are of width m degrees, and the control signalis phase-shifted in a range between m degrees and 360−m+360·n degrees ofa frequency of the input signal (no overlap of pulses between controlsignal and phase-shifted control signal). In embodiments, the controlsignal is phase shifted to a phase-shifted control signal in order toadjust at least one frequency of the re-radiated signal. In furtherembodiments, the control signal is phase shifted to a phase-shiftedcontrol signal in order to adjust at least one frequency of there-radiated signal to be above a frequency range of interest of theinput signal.

[0702] In step 11706, the input signal is frequency down-converted witha second universal frequency down-conversion module to a seconddown-converted signal, wherein the input signal is down-convertedaccording to the phase-shifted control signal. The second universalfrequency down-conversion module is used as a placebo universalfrequency down-conversion module.

[0703] In step 11708, a signal is re-radiated that is at least afunction of the control signal and the phase-shifted control signal.

[0704] Exemplary receiver with placebo circuit embodiments are describedabove. However, it should be understood that these examples are providedfor illustrative purposes only. The invention is not limited to theseembodiments. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. For example, further placebo UFD moduleswith additional placebo control signals may be added. The invention isintended and adapted to include such alternate embodiments.

[0705] 7.2.2.3 Reducing Re-Radiation with Adjacent Apertures

[0706] Potential control signal circuit re-radiation may be reduced oreliminated by the use of adjacent control signal pulses or apertures. Bycreating control signal pulses that are adjacent, the rising and fallingedges of adjacent pulses may partially or entirely cancel out anyre-radiation due to the individual pulses.

[0707]FIG. 102 illustrates an adjacent apertures receiver circuit 10200,according to an embodiment of the present invention. Adjacent aperturesreceiver circuit 10200 comprises a first UFD module 10202, a second UFDmodule 10204, a control signal generator 10206, and a phase shifter10208.

[0708] An input RF signal 10214 is received by first UFD module 10202.First UFD module 10202 down-converts input RF signal 10214 according toa control signal 10220, which is output by control signal generator10206, in a manner as described elsewhere herein. First UFD module 10202outputs first output signal 10216. First UFD module 10202 comprises afirst UFT module 10210.

[0709] Phase shifter 10208 receives control signal 10220, and outputs aphase-shifted control signal 10222. In an embodiment, the width ofpulses on control signal 10220 and on phase-shifted control signal 10222approach π radians, although other values could be used. Phase-shiftedcontrol signal 10222 is preferably shifted by π radians from controlsignal 10220, although other values could be used.

[0710] Input RF signal 10214 is received by second UFD module 10204.Second UFD module 10204 down-converts input RF signal 10214 according tophase-shifted control signal 10222, in a manner as described elsewhereherein. Second UFD module 10204 outputs second output signal 10218.Second UFD module 10204 comprises a second UFT module 10212.

[0711]FIG. 103 shows an exemplary control signal waveform 10302, and acorresponding π-shifted control signal waveform 10304. The potentiallyre-radiated signals (and their associated frequency spectrums) due tocontrol signal 10220 and phase-shifted control signal 10222 will berelated to their waveforms and frequency spectrums, which may berepresented by control signal waveform 10302 and π-shifted controlsignal waveform 10304, respectively, for example.

[0712] Because the width of pulses on control signal waveform 10302 andπ-shifted control signal waveform 10304 are equal to or less than πradians, their combined potentially re-radiated signal will be relatedto combined signal waveform 10306. As control signal waveform 10302 andn-shifted control signal waveform 10306 approach having pulse widthsequal to π radians, combined signal waveform 10306 will approach theequivalent of a DC level, with a voltage level substantially equivalentto the pulse amplitudes. In other words, combined signal waveform 10306will approach a DC level because as pulses of waveforms 10302 and 10304approach a width of π, the rising and falling edges of the waveforms10302 and 10304 will increasingly cancel each other.

[0713] The use of adjacent apertures may lead to reduced levels ofcircuit re-radiation, and improved circuit performance. Re-radiatedsignal components will be due to combined signal waveform 10306.Specifically, re-radiated signal components will be due to transitionsfrom low to high and high to low in waveform 10306, shown as spikes10308, but the frequency content of such re-radiated signal componentsdue to spikes 10308 will primarily be above the frequency bands ofinterest.

[0714]FIG. 118A depicts a flowchart 11800 that illustrates operationalsteps corresponding to the structure of FIG. 102 and waveforms of FIG.103, for down-converting an input signal and altering circuitre-radiation, according to an embodiment of the present invention. Theinvention is not limited to this operational description. Rather, itwill be apparent to persons skilled in the relevant art(s) from theteachings herein that other operational control flows are within thescope and spirit of the present invention. In the following discussion,the steps in FIG. 118A will be described.

[0715] In step 11802, an input signal is frequency down-converted with afirst universal frequency down-conversion module to a firstdown-converted signal, wherein the input signal is down-convertedaccording to a control signal, wherein the control signal comprises atrain of pulses, wherein the pulses have widths less than or equal to180+360·n degrees of a frequency of the input signal, wherein n is anyinteger greater than or equal to 0.

[0716] In step 11804, the control signal is phase shifted, wherein thecontrol signal is phase-shifted by 180+360·n degrees of a frequency ofthe input signal. In an embodiment, the pulses of the control signal aresubstantially adjacent to pulses of the phase-shifted control signal.

[0717] In step 11806, the input signal is frequency down-converted witha second universal frequency down-conversion module to a seconddown-converted signal, wherein the input signal is down-convertedaccording to the phase-shifted control signal.

[0718] In step 11808, a signal is re-radiated that is at least afunction of the control signal and the phase-shifted control signal. Inan embodiment, a spike is formed in the re-radiated signal at atransition of the adjacent pulses of the control signal and thephase-shifted control signal. In an embodiment, a voltage amplitude ofthe spike approaches zero as the pulses of the control signal and thepulses of the phase-shifted control signal approach 180+360·n degrees inwidth (i.e., the pulses become more adjacent). In an embodiment, atleast one frequency of the spike is above a frequency range of interestof the input signal. In embodiments, as the pulse widths approach180+360·n degrees of a frequency of the input signal, the re-radiatedsignal approaches a DC level.

[0719] In alternate embodiments, other adjacent control signal pulseconfigurations may be used. FIG. 104 illustrates an exemplary adjacentapertures receiver circuit 10400, according to an embodiment of thepresent invention. Circuit 10400 operates substantially similarly tocircuit 10200, but produces four control signals instead of two. Controlsignal generator 10410 outputs control signal 10432, with pulse widthsof π, which occur once every 4π radians. Circuit 10400 comprises a firstphase shifter 10412, a second phase shifter 10440, and a third phaseshifter 10442, each of which further shifts the phase of control signal10432 by π radians. In this manner, four adjacent control signalgenerator pulses, each of pulse width π radians, are generated thatapproach the equivalent of a DC level. Other control signal aperturedurations and/or sequences will be apparent to persons skilled in therelevant art(s) based on the teachings contained herein.

[0720] Additional control signals may be used to produce even longerstrings of adjacent pulses.

[0721] Furthermore, the use of adjacent apertures may reduce the needfor input impedance matching and tank circuitry. This is because withadjacent apertures, the UFT modules in combination are closed for longerfractions of a control signal cycle and hence, the input signal is beingstored more continuously (by a storage module, for example). Because theinput signal is being stored more continuously, there is lessopportunity or need to store the input signal in one or more input tankcircuits during the periods when the UFT modules are open. In otherwords, more of the energy of the input waveform is being stored and usedwith adjacent apertures. Furthermore, having the UFT module(s) closedfor longer periods of time affects the circuit input impedance, and mayalter or decrease the need for input impedance matching.

[0722]FIG. 118B depicts a continuation of flowchart 11800 thatillustrates additional operational steps to those shown in FIG. 118Acorresponding to further adjacent aperture generators, such as shown inFIG. 104, for down-converting an input signal and altering circuitre-radiation, according to an embodiment of the present invention. Theinvention is not limited to this operational description. Rather, itwill be apparent to persons skilled in the relevant art(s) from theteachings herein that other operational control flows are within thescope and spirit of the present invention. In the following discussion,the steps in FIG. 118B will be described.

[0723] In step 11810, a phase-shifted control signal is phase shifted toa further phase-shifted control signal. The phase-control signal isphase shifted by the same amount as the prior phase shifter. This causesthe current aperture or pulse to be the same width as, and adjacent to,the prior aperture.

[0724] In step 11812, the input signal is frequency down-converted witha further universal frequency down-conversion module to a correspondingdown-converted signal, wherein the input signal is down-convertedaccording to the further phase-shifted control signal.

[0725] In step 11814, a signal is re-radiated that is a function of atleast the control signal and the phase-shifted control signals.

[0726] In step 11816, operation proceeds to step 11810 if the number ofuniversal frequency down-conversion modules (adjacent apertures) is lessthan some desired number x. This process forms a chain of adjacentapertures, of a number of pulses x.

[0727] Exemplary receivers using adjacent apertures embodiments aredescribed above. However, it should be understood that these examplesare provided for illustrative purposes only. The invention is notlimited to these embodiments. Alternate embodiments (includingequivalents, extensions, variations, deviations, etc., of theembodiments described herein) will be apparent to persons skilled in therelevant art(s) based on the teachings contained herein. For example,control signals with pulses of widths other than π radians where thecontrol signals have different pulse widths, may be used. The inventionis intended and adapted to include such alternate embodiments.

[0728] 7.2.3 Additional DC Offset and Re-Radiation Reduction Embodiments

[0729] Exemplary embodiments for DC Offset and/or re-radiation reductionor cancellation are described above. Such embodiments may be used aloneor in combination, based on the application and on implementationissues. It should be understood that these examples are provided forillustrative purposes only. The invention is not limited to theseembodiments. Alternate embodiments (including equivalents, extensions,variations, deviations, etc., of the embodiments described herein) willbe apparent to persons skilled in the relevant art(s) based on theteachings contained herein. For example, many of the componentsdescribed herein are optional, whether or not explicitly indicated assuch. The invention is intended and adapted to include such alternateembodiments.

[0730] 7.3 Example Embodiments to Improve Dynamic Range

[0731] Receivers, amplifiers, and other electronic circuits, may sufferfrom problems related to dynamic range. Generally, “dynamic range”refers to the ratio of the maximum to minimum signal input capabilityover which an amplifier or other component can operate within somespecified range of performance. For instance, if an input signal to anamplifier causes the amplifier to exceed its dynamic range, i.e., theinput signal amplitude is too large, the amplifier may no longer amplifyproperly, the amplifier may rail, and/or may operate in a non-linearregion. In a receiver, the signal being amplified may be adown-converted signal. If the dynamic range of the amplifier, or othercomponent, is exceeded, the value of the down-converted signal may beadversely affected.

[0732] The concept of dynamic range is further described in thefollowing sub-sections. Furthermore, example methods and systems areprovided in subsequent sections below for improving dynamic range.

[0733] 7.3.1 Adjusting Universal Frequency Down-Conversion ModuleDynamic Range

[0734] Some circuit implementations may suffer from a lack of dynamicrange. For instance, when input RF signals become too high or too low,they may cause a switch in a UFT module to remain continuously open orclosed, regardless of the level of the control signal. This may resultin problems with output signal linearity and output signal clipping,which may lead to errors in decoding the baseband output signal.

[0735]FIG. 105 illustrates an exemplary circuit for improving dynamicrange, according to an embodiment of the present invention. Improveddynamic range circuit 10500 of FIG. 105 comprises a impedance match10502, a tank circuit 10504, a UFD module 10506, and a bias circuit10508. Bias circuit 10508 is used to adjust the center point of theinput voltage range for UFD module 10506, providing for greater inputsignal range.

[0736] In an embodiment, impedance match 10502 comprises an inductor10510. The operation of the present and of additional embodiments forimpedance match 10502 are further described elsewhere herein.

[0737] In an embodiment, tank circuit 10504 comprises a capacitor 10512and an inductor 10514. The operation of the present and of additionalembodiments for tank circuit 10504 are further described elsewhereherein.

[0738] UFD module 10506 comprises a UFT module 10516, a storage module10520, and a first voltage reference 10524. The operation of the presentand of additional embodiments for UFD module 10506 are further describedelsewhere herein. UFT module 10516 comprises a MOSFET switch 10518 inthe example embodiment of FIG. 105. Storage module 10520 comprises acapacitor 10522 in the example embodiment of FIG. 105. The structure andoperation of the present and of additional embodiments for UFT module10516 and storage module 10520 are further described elsewhere herein.

[0739] An input RF signal 10540 is input through impedance match 10502to be received by a first terminal 10550 of UFD module 10506. Firstterminal 10550 of UFD module 10506 is coupled to a first terminal 10552of tank circuit 10504. MOSFET switch 10518 in UFD module 10506down-converts input RF signal 10540 according to a control signal 10548,which is output by control signal generator 10526. The output of MOSFETswitch 10518 is stored in storage module 10520. MOSFET switch 10518outputs an output signal 10542. In the example embodiment of FIG. 105,MOSFET switch 10518 comprises a first terminal 10564 coupled to firstterminal 10552 of tank circuit 10504, a second terminal 10566 coupled tooutput signal 10542, and a third terminal 10568 (gate) coupled tocontrol signal 10548.

[0740] Control signal generator 10526 generates control signal 10548, asdescribed elsewhere herein. Control signal 10548 preferably comprises aperiodic signal, which preferably comprises a string of pulses. Thesepulses vary between a minimum and maximum voltage. For example, controlsignal 10548 may output pulses that vary between 0 volts and 2 volts, asshown in FIG. 106A.

[0741] Input RF signal 10540 also comprises a range of signal values.For instance, input RF signal 10540 may vary between +0.75 volts and−0.75 volts, as shown in FIG. 106B. In the current example, when thevalue of input RF signal 10540 is equal to −0.75 volts, this value isless than the minimum voltage of control signal 10548 (0 volts) appliedto MOSFET switch 10518, and hence MOSFET switch 10518 will be in theclosed state for all values of control signal 10548 because the voltagefrom terminal 10568 (gate) to terminal 10564 of MOSFET switch 10518 isalways positive, causing MOSFET switch 10518 to always conduct.

[0742] Likewise, it will be recognized by persons skilled in therelevant art(s) that input RF signal 10540 may comprise signalamplitudes greater than the maximum voltage of control signal 10548 (notillustrated in FIGS. 106A and 106B) applied to MOSFET switch 10518. Wheninput RF signal 10540 is equal to such a value, MOSFET switch 10518 willremain in the open state, for all values of control signal 10548,because the voltage from terminal 10568 (gate) to terminal 10564 ofMOSFET switch 10518 would always be negative, preventing MOSFET switch10518 from ever conducting. Both conditions where control signal 10548cannot effect switching of MOSFET switch 10518 are undesirable.

[0743] One solution is to modify the voltage swing of control signal10548 such that it varies from +0.75 to −0.75 volts or greater, as doesinput RF signal 10540. This solution may not be possible in allsituations, however. For instance, this solution may not be possiblewhen only a single voltage supply is available.

[0744] A further solution for this problem is to bias input RF signal10540 such that it varies within the maximum and minimum voltage rangeof pulses of control signal 10548. Thus, as long as input RF signal10540 varies within the voltage range of control signal 10548, controlsignal 10548 will control the turning on and turning off of MOSFETswitch 10518. FIG. 106C shows an example biased input RF signal 10544,that is biased to vary between +1.75 and +0.25 volts, within the rangeof control signal 10548.

[0745] Bias circuit 10508 is used to adjust the bias applied to input RFsignal 10540. (It is noted that other bias configurations couldalternatively be used.) Bias circuit 10508 comprises a second voltagereference 10528, a first resistor 10530, an optional capacitor 10532, athird voltage reference 10534, a second resistor 10536, and a fourthvoltage reference 10538.

[0746] A first terminal 10554 of first resistor 10530 is coupled to afirst voltage reference 10528. A second terminal 10556 of first resistor10530 is coupled to a first terminal 10558 of second resistor 10536 tocreate a bias point 10546. Bias point 10546 is coupled to a secondterminal 10560 of tank circuit 10504. A second terminal 10562 of secondresistor 10536 is coupled to fourth voltage reference 10538.

[0747] First resistor 10530 and second resistor 10536 form a voltagedivider circuit, to create bias point 10546, as would be understood bypersons skilled in the relevant art(s) from the teachings herein. Biaspoint 10546 provides a biasing voltage for input RF signal 10540. Abiased input RF signal 10544 is equal to input RF signal 10540 adjusted(e.g., added or subtracted) by the amount of voltage at bias point10546. In a preferred embodiment, biased input RF signal 10544 may bebiased at the midpoint of the voltage swing of control signal 10548. Forexample, biased input RF signal 10544 may be biased by bias point 10546with a level of one volt, for a 0 volt to 2 volt varying control signal10548.

[0748] Optional capacitor 10532 coupled between bias point 10546 andthird voltage reference 10534 may be optionally inserted to aid instabilizing bias point 10546.

[0749] Other embodiments for bias circuit 10508 will be apparent topersons skilled in the relevant art(s) from the teachings herein. Forinstance, FIG. 107 illustrates an exemplary bias circuit 10708 accordingto an embodiment of the present invention, wherein a tank circuit 10504and/or an impedance match circuit 10502 as shown in FIG. 105 are notpresent. Bias circuit 10708 in FIG. 107 adjusts a bias point for aninput to UFD module 10506. Bias circuit 10708 of FIG. 107 places a biasdirectly on input RF signal 10540, as opposed to bias circuit 10508 ofFIG. 105 which applies a bias voltage through tank circuit 10504.

[0750]FIG. 119 depicts a flowchart 11900 that illustrates operationalsteps corresponding to FIGS. 105-107, for improving dynamic range,according to an embodiment of the present invention. The invention isnot limited to this operational description. Rather, it will be apparentto persons skilled in the relevant art(s) from the teachings herein thatother operational control flows are within the scope and spirit of thepresent invention. In the following discussion, the steps in FIG. 119will be described.

[0751] In step 11902, a bias voltage is applied to an input signal. Inembodiments, the center voltage of the input signal is adjusted byapplication of the bias voltage. In embodiments, the input signal iscoupled to a center terminal of a resistor divider circuit, whichsupplies the bias voltage. In an embodiment, a tank circuit is used tocouple the input signal to the center terminal of the resistor dividercircuit.

[0752] In step 11904, the biased input signal is frequencydown-converted with a first universal frequency down-conversion moduleto a down-converted signal.

[0753] Other embodiments for improving dynamic range include the use ofcomplementary FETs. Complementary FET embodiments are further describedin the co-pending U.S. Patent Application entitled “Method and Systemfor Down-converting Electromagnetic Signals Having Optimized SwitchStructures,” Ser. No. 09/293,095. Complementary FETs also have theadvantage of using control signals of opposite polarity, which tends toreduce or cancel re-radiation due to a control signal.

[0754] Other circuit embodiments for improving dynamic range includemodifying control signal pulse amplitude, and/or modifying the switch,or FET, size, as would be understood by persons skilled in the relevantart(s) from the teachings herein. It should be understood that the abovebias circuit examples are provided for illustrative purposes only. Theinvention is not limited to these embodiments. Alternate embodiments(including equivalents, extensions, variations, deviations, etc., of theembodiments described herein) will be apparent to persons skilled in therelevant art(s) based on the teachings contained herein. The inventionis intended and adapted to include such alternate embodiments.

[0755] 7.4 Example Receiver and Transmitter Embodiments for AddressingDC Offset and Re-Radiation

[0756] In this section, embodiments, according to the present invention,are provided for reducing or eliminating DC offset and/or reducing oreliminating circuit re-radiation in receivers, including I/Q modulationreceivers and other modulation scheme receivers. These embodiments aredescribed herein for purposes of illustration, and not limitation. Theinvention is not limited to these embodiments. Alternate embodiments(including equivalents, extensions, variations, deviations, etc., of theembodiments described herein) will be apparent to persons skilled in therelevant art(s) based on the teachings contained herein. The inventionis intended and adapted to include such alternate embodiments.

[0757] 7.4.1 Example I/Q Modulation Receiver Embodiments

[0758]FIG. 25 illustrates an exemplary I/Q modulation receiver 2500,according to an embodiment of the present invention. I/Q modulationreceiver 2500 has additional advantages of reducing or eliminatingunwanted DC offsets and circuit re-radiation.

[0759] I/Q modulation receiver 2500 comprises a first UFD module 2502, afirst optional filter 2504, a second UFD module 2506, a second optionalfilter 2508, a third UFD module 2510, a third optional filter 2512, afourth UFD module 2514, a fourth filter 2516, an optional LNA 2518, afirst differential amplifier 2520, a second differential amplifier 2522,and an antenna 2572.

[0760] I/Q modulation receiver 2500 receives, down-converts, anddemodulates a I/Q modulated RF input signal 2582 to an I baseband outputsignal 2584, and a Q baseband output signal 2586. I/Q modulated RF inputsignal comprises a first information signal and a second informationsignal that are I/Q modulated onto an RF carrier signal. I basebandoutput signal 2584 comprises the first baseband information signal. Qbaseband output signal 2586 comprises the second baseband informationsignal.

[0761] Antenna 2572 receives I/Q modulated RF input signal 2582. I/Qmodulated RF input signal 2582 is output by antenna 2572 and received byoptional LNA 2518. When present, LNA 2518 amplifies I/Q modulated RFinput signal 2582, and outputs amplified I/Q signal 2588.

[0762] First UFD module 2502 receives amplified I/Q signal 2588. FirstUFD module 2502 down-converts the I-phase signal portion of amplifiedinput I/Q signal 2588 according to an I control signal 2590. First UFDmodule 2502 outputs an I output signal 2598.

[0763] In an embodiment, first UFD module 2502 comprises a first storagemodule 2524, a first UFT module 2526, and a first voltage reference2528. In an embodiment, a switch contained within first UFT module 2526opens and closes as a function of I control signal 2590. As a result ofthe opening and closing of this switch, which respectively couples andde-couples first storage module 2524 to and from first voltage reference2528, a down-converted signal, referred to as I output signal 2598,results. First voltage reference 2528 may be any reference voltage, andis preferably ground. I output signal 2598 is stored by first storagemodule 2524.

[0764] In a preferred embodiment, first storage module 2524 comprises afirst capacitor 2574. In addition to storing I output signal 2598, firstcapacitor 2574 reduces or prevents a DC offset voltage resulting fromabove described charge injection from appearing on I output signal 2598,in a similar fashion to that of capacitor 9126 shown in FIG. 91. Referto section 7.2.1.3 above for further discussion on reducing oreliminating charge injection with a series capacitor such as capacitor9126.

[0765] I output signal 2598 is received by optional first filter 2504.When present, first filter 2504 is a high pass filter to at least filterI output signal 2598 to remove any carrier signal “bleed through”. In apreferred embodiment, when present, first filter 2504 comprises a firstresistor 2530, a first filter capacitor 2532, and a first filter voltagereference 2534. Preferably, first resistor 2530 is coupled between Ioutput signal 2598 and a filtered I output signal 2507, and first filtercapacitor 2532 is coupled between filtered I output signal 2507 andfirst filter voltage reference 2534. Alternately, first filter 2504 maycomprise any other applicable filter configuration as would beunderstood by persons skilled in the relevant art(s). First filter 2504outputs filtered I output signal 2507.

[0766] Second UFD module 2506 receives amplified I/Q signal 2588. SecondUFD module 2506 down-converts the inverted I-phase signal portion ofamplified input I/Q signal 2588 according to an inverted I controlsignal 2592. Second UFD module 2506 outputs an inverted I output signal2501.

[0767] In an embodiment, second UFD module 2506 comprises a secondstorage module 2536, a second UFT module 2538, and a second voltagereference 2540. In an embodiment, a switch contained within second UFTmodule 2538 opens and closes as a function of inverted I control signal2592. As a result of the opening and closing of this switch, whichrespectively couples and de-couples second storage module 2536 to andfrom second voltage reference 2540, a down-converted signal, referred toas inverted I output signal 2501, results. Second voltage reference 2540may be any reference voltage, and is preferably ground. Inverted Ioutput signal 2501 is stored by second storage module 2536.

[0768] In apreferred embodiment, second storage module 2536 comprises asecond capacitor 2576. In addition to storing inverted I output signal2501, second capacitor 2576 reduces or prevents a DC offset voltageresulting from above described charge injection from appearing oninverted I output signal 2501, in a similar fashion to that of capacitor9126 shown in FIG. 91. Refer to section 7.2.1.3 above for furtherdiscussion on reducing or eliminating charge injection with a seriescapacitor such as capacitor 9126.

[0769] Inverted I output signal 2501 is received by optional secondfilter 2508. When present, second filter 2508 is a high pass filter toat least filter inverted I output signal 2501 to remove any carriersignal “bleed through”. In a preferred embodiment, when present, secondfilter 2508 comprises a second resistor 2542, a second filter capacitor2544, and a second filter voltage reference 2546. Preferably, secondresistor 2542 is coupled between inverted I output signal 2501 and afiltered inverted I output signal 2509, and second filter capacitor 2544is coupled between filtered inverted I output signal 2509 and secondfilter voltage reference 2546. Alternately, second filter 2508 maycomprise any other applicable filter configuration as would beunderstood by persons skilled in the relevant art(s). Second filter 2508outputs filtered inverted I output signal 2509.

[0770] First differential amplifier 2520 receives filtered I outputsignal 2507 at its non-inverting input and receives filtered inverted Ioutput signal 2509 at its inverting input. First differential amplifier2520 subtracts filtered inverted I output signal 2509 from filtered Ioutput signal 2507, amplifies the result, and outputs I baseband outputsignal 2584. Other suitable subtractor and/or amplification modules maybe substituted for first differential amplifier 2520, and seconddifferential amplifier 2522, as would be understood by persons skilledin the relevant art(s) from the teachings herein. Because filteredinverted I output signal 2509 is substantially equal to an invertedversion of filtered I output signal 2507, 1 baseband output signal 2584is substantially equal to filtered I output signal 2509, with itsamplitude doubled. Furthermore, filtered I output signal 2507 andfiltered inverted I output signal 2509 may comprise substantially equalnoise and DC offset contributions of the same polarity from priordown-conversion circuitry, including first UFD module 2502 and secondUFD module 2506, respectively. When first differential amplifier 2520subtracts filtered inverted I output signal 2509 from filtered I outputsignal 2507, these noise and DC offset contributions substantiallycancel each other.

[0771] Third UFD module 2510 receives amplified I/Q signal 2588. ThirdUFD module 2510 down-converts the Q-phase signal portion of amplifiedinput I/Q signal 2588 according to an Q control signal 2594. Third UFDmodule 2510 outputs an Q output signal 2503.

[0772] In an embodiment, third UFD module 2510 comprises a third storagemodule 2548, a third UFT module 2550, and a third voltage reference2552. In an embodiment, a switch contained within third UFT module 2550opens and closes as a function of Q control signal 2594. As a result ofthe opening and closing of this switch, which respectively couples andde-couples third storage module 2548 to and from third voltage reference2552, a down-converted signal, referred to as Q output signal 2503,results. Third voltage reference 2552 may be any reference voltage, andis preferably ground. Q output signal 2503 is stored by third storagemodule 2548.

[0773] In a preferred embodiment, third storage module 2548 comprises athird capacitor 2578. In addition to storing Q output signal 2503, thirdcapacitor 2578 reduces or prevents a DC offset voltage resulting fromabove described charge injection from appearing on Q output signal 2503,in a similar fashion to that of capacitor 9126 shown in FIG. 91. Referto section 7.2.1.3 above for further discussion on reducing oreliminating charge injection with a series capacitor such as capacitor9126.

[0774] Q output signal 2503 is received by optional third filter 2512.When present, third filter 2512 is a high pass filter to at least filterQ output signal 2503 to remove any carrier signal “bleed through”. In apreferred embodiment, when present, third filter 2512 comprises a thirdresistor 2554, a third filter capacitor 2558, and a third filter voltagereference 2558. Preferably, third resistor 2554 is coupled between Qoutput signal 2503 and a filtered Q output signal 2511, and third filtercapacitor 2556 is coupled between filtered Q output signal 2511 andthird filter voltage reference 2558. Alternately, third filter 2512 maycomprise any other applicable filter configuration as would beunderstood by persons skilled in the relevant art(s). Third filter 2512outputs filtered Q output signal 2511.

[0775] Fourth UFD module 2514 receives amplified I/Q signal 2588. FourthUFD module 2514 down-converts the inverted Q-phase signal portion ofamplified input I/Q signal 2588 according to an inverted Q controlsignal 2596. Fourth UFD module 2514 outputs an inverted Q output signal2505.

[0776] In an embodiment, fourth UFD module 2514 comprises a fourthstorage module 2560, a fourth UFT module 2562, and a fourth voltagereference 2564. In an embodiment, a switch contained within fourth UFTmodule 2562 opens and closes as a function of inverted Q control signal2596. As a result of the opening and closing of this switch, whichrespectively couples and de-couples fourth storage module 2560 to andfrom fourth voltage reference 2564, a down-converted signal, referred toas inverted Q output signal 2505, results. Fourth voltage reference 2564may be any reference voltage, and is preferably ground. Inverted Qoutput signal 2505 is stored by fourth storage module 2560.

[0777] In a preferred embodiment, fourth storage module 2560 comprises afourth capacitor 2580. In addition to storing inverted Q output signal2505, fourth capacitor 2580 reduces or prevents a DC offset voltageresulting from above described charge injection from appearing oninverted Q output signal 2505, in a similar fashion to that of capacitor9126 shown in FIG. 91. Refer to section 7.2.1.3 above for furtherdiscussion on reducing or eliminating charge injection with a seriescapacitor such as capacitor 9126.

[0778] Inverted Q output signal 2505 is received by optional fourthfilter 2516. When present, fourth filter 2516 is a high pass filter toat least filter inverted Q output signal 2505 to remove any carriersignal “bleed through”. In a preferred embodiment, when present, fourthfilter 2516 comprises a fourth resistor 2566, a fourth filter capacitor2568, and a fourth filter voltage reference 2570. Preferably, fourthresistor 2566 is coupled between inverted Q output signal 2505 and afiltered inverted Q output signal 2513, and fourth filter capacitor 2568is coupled between filtered inverted Q output signal 2513 and fourthfilter voltage reference 2570. Alternately, fourth filter 2516 maycomprise any other applicable filter configuration as would beunderstood by persons skilled in the relevant art(s). Fourth filter 2516outputs filtered inverted Q output signal 2513.

[0779] Second differential amplifier 2522 receives filtered Q outputsignal 2511 at its non-inverting input and receives filtered inverted Qoutput signal 2513 at its inverting input. Second differential amplifier2522 subtracts filtered inverted Q output signal 2513 from filtered Qoutput signal 2511, amplifies the result, and outputs Q baseband outputsignal 2586. Because filtered inverted Q output signal 2513 issubstantially equal to an inverted version of filtered Q output signal2511, Q baseband output signal 2586 is substantially equal to filtered Qoutput signal 2513, with its amplitude doubled. Furthermore, filtered Qoutput signal 2511 and filtered inverted Q output signal 2513 maycomprise substantially equal noise and DC offset contributions of thesame polarity from prior down-conversion circuitry, including third UFDmodule 2510 and fourth UFD module 2514, respectively. When seconddifferential amplifier 2522 subtracts filtered inverted Q output signal2513 from filtered Q output signal 2511, these noise and DC offsetcontributions substantially cancel each other.

[0780]FIG. 120 depicts a flowchart 12000 that illustrates operationalsteps corresponding to FIG. 25, for down-converting a RF I/Q modulatedsignal and reducing DC offset voltages, according to an embodiment ofthe present invention. The invention is not limited to this operationaldescription. Rather, it will be apparent to persons skilled in therelevant art(s) from the teachings herein that other operational controlflows are within the scope and spirit of the present invention. In thefollowing discussion, the steps in FIG. 120 will be described.

[0781] In step 12002, an input signal is received, wherein the inputsignal comprises an RF I/Q modulated signal.

[0782] In step 12004, the input signal is frequency down-converted witha first universal frequency down-conversion module to a firstdown-converted signal, according to a first control signal. In anembodiment, the input signal is frequency down-converted to anon-inverted I-phase signal portion of the RF I/Q modulated signal. Forinstance, in an embodiment, a first phase of the in-phase signal portionof the RF I/Q modulated signal is under-sampled. In an embodiment, theRF I/Q modulated signal may be under-sampled every 3.0 cycles of afrequency of the RF I/Q modulated signal by the first control signal.Furthermore, in embodiments, a first DC offset voltage in the firstdown-converted signal is reduced by a capacitor of the first universalfrequency down-conversion module.

[0783] In step 12006, the input signal is frequency down-converted witha second universal frequency down-conversion module to a seconddown-converted signal, according to a second control signal. In anembodiment, the input signal is frequency down-converted to an invertedQ-phase signal portion of the RF I/Q modulated signal. For instance, inan embodiment, a second phase of the in-phase signal portion of the RFI/Q modulated signal is under-sampled, wherein the second phase of thein-phase signal portion is of an opposite phase to the first phaseunder-sampled of the in-phase signal portion. The RF I/Q modulatedsignal may be sampled 1.5 cycles of a frequency of the RF I/Q modulatedsignal after under-sampling the RF I/Q modulated signal in step 12004,for example. Furthermore, in embodiments, a second DC offset voltage inthe second down-converted signal is reduced by a capacitor of the seconduniversal frequency down-conversion module.

[0784] In step 12008, the second down-converted signal is subtractedfrom the first down-converted signal to form a first output signal. Inembodiments, a first DC offset voltage in the first down-convertedsignal and a second DC offset voltage in the second down-convertedsignal cancel one another.

[0785] In step 12010, the input signal is frequency down-converted witha third universal frequency down-conversion module to a thirddown-converted signal, according to a third control signal. In anembodiment, the input signal is frequency down-converted to anon-inverted Q-phase signal portion of the RF I/Q modulated signal. Forinstance, in an embodiment, a third phase of the quadrature-phase signalportion of the RF I/Q modulated signal is under-sampled. The RF I/Qmodulated signal may be under-sampled 0.75 cycles of the frequency ofthe RF I/Q modulated signal after under-sampling of the RF I/Q modulatedsignal occurs in step 12004, for example. Furthermore, in embodiments, athird DC offset voltage in the third down-converted signal is reduced bya capacitor of the third universal frequency down-conversion module.

[0786] In step 12012, the input signal is frequency down-converted witha fourth universal frequency down-conversion module to a fourthdown-converted signal, according to a fourth control signal. In anembodiment, the input signal is frequency down-converted to an invertedI-phase signal portion of the RF I/Q modulated signal. For instance, inan embodiment, a fourth phase of the quadrature-phase signal portion ofthe RF I/Q modulated signal is under-sampled, wherein the fourth phaseof the quadrature-phase signal portion is of an opposite phase to thethird phase under-sampled of the quadrature-phase signal portion. In anembodiment, the RF I/Q modulated signal may be sampled 1.5 cycles of thefrequency of the RF I/Q modulated signal after under-sampling of the RFI/Q modulated signal occurs in step 12004, for example. Furthermore, inembodiments, a fourth DC offset voltage in the fourth down-convertedsignal is reduced by a capacitor of fourth universal frequencydown-conversion module.

[0787] In step 12014, the fourth down-converted signal is subtractedfrom the third down-converted signal to form a second output signal. Inembodiments, a third DC offset voltage in the third down-convertedsignal and a fourth DC offset voltage in the fourth down-convertedsignal cancel one another.

[0788] In step 12016, a signal is re-radiated that comprises attenuatedcomponents of first, second, third, and fourth control signal pulses,wherein the attenuated components of the first, second, third, andfourth control signal pulses form a cumulative frequency, as discussedabove.

[0789] In step 12018, the first, second, third, and fourth controlsignal pulses are configured such that the cumulative frequency isgreater than a frequency of the input signal, as discussed above.

[0790] 7.4.1.1 Example I/Q Modulation Control Signal GeneratorEmbodiments

[0791]FIG. 26 illustrates an exemplary block diagram for I/Q modulationcontrol signal generator 2600, according to an embodiment of the presentinvention. I/Q modulation control signal generator 2600 generates Icontrol signal 2590, inverted I control signal 2592, Q control signal2594, and inverted Q control signal 2596 used by I/Q modulation receiver2500 of FIG. 25. I control signal 2590 and inverted I control signal2592 operate to down-convert the I-phase portion of an input I/Qmodulated RF signal. Q control signal 2594 and inverted Q control signal2596 act to down-convert the Q-phase portion of the input I/Q modulatedRF signal. Furthermore, I/Q modulation control signal generator 2600 hasthe advantage of generating control signals in a manner such thatresulting collective circuit re-radiation is radiated at one or morefrequencies outside of the frequency range of interest. For instance,potential circuit re-radiation is radiated at a frequency substantiallygreater than that of the input RF carrier signal frequency.

[0792] I/Q modulation control signal generator 2600 comprises a localoscillator 2602, a first divide-by-two module 2604, a 180 degree phaseshifter 2606, a second divide-by-two module 2608, a first pulsegenerator 2610, a second pulse generator 2612, a third pulse generator2614, and a fourth pulse generator 2616.

[0793] Local oscillator 2602 outputs an oscillating signal 2618. FIG. 27shows an exemplary oscillating signal 2618.

[0794] First divide-by-two module 2604 receives oscillating signal 2618,divides oscillating signal 2618 by two, and outputs a half frequency LOsignal 2620 and a half frequency inverted LO signal 2626. FIG. 27 showsan exemplary half frequency LO signal 2620. Half frequency inverted LOsignal 2626 is an inverted version of half frequency LO signal 2620.First divide-by-two module 2604 may be implemented in circuit logic,hardware, software, or any combination thereof, as would be known bypersons skilled in the relevant art(s). 180 degree phase shifter 2606receives oscillating signal 2618, shifts the phase of oscillating signal2618 by 180 degrees, and outputs phase-shifted LO signal 2622. 180degree phase shifter 2606 may be implemented in circuit logic, hardware,software, or any combination thereof, as would be known by personsskilled in the relevant art(s). In alternative embodiments, otheramounts of phase shift may be used.

[0795] Second divide-by two module 2608 receives phase-shifted LO signal2622, divides phase-shifted LO signal 2622 by two, and outputs a halffrequency phase-shifted LO signal 2624 and a half frequency invertedphase-shifted LO signal 2628. FIG. 27 shows an exemplary half frequencyphase-shifted LO signal 2624. Half frequency inverted phase-shifted LOsignal 2628 is an inverted version of half frequency phase-shifted LOsignal 2624. Second divide-by-two module 2608 may be implemented incircuit logic, hardware, software, or any combination thereof, as wouldbe known by persons skilled in the relevant art(s).

[0796] First pulse generator 2610 receives half frequency LO signal2620, generates an output pulse whenever arising edge is received onhalf frequency LO signal 2620, and outputs I control signal 2590. FIG.27 shows an exemplary I control signal 2590.

[0797] Second pulse generator 2612 receives half frequency inverted LOsignal 2626, generates an output pulse whenever a rising edge isreceived on half frequency inverted LO signal 2626, and outputs invertedI control signal 2592. FIG. 27 shows an exemplary inverted I controlsignal 2592.

[0798] Third pulse generator 2614 receives half frequency phase-shiftedLO signal 2624, generates an output pulse whenever a rising edge isreceived on half frequency phase-shifted LO signal 2624, and outputs Qcontrol signal 2594. FIG. 27 shows an exemplary Q control signal 2594.

[0799] Fourth pulse generator 2616 receives half frequency invertedphase-shifted LO signal 2628, generates an output pulse whenever arising edge is received on half frequency inverted phase-shifted LOsignal 2628, and outputs inverted Q control signal 2596. FIG. 27 showsan exemplary inverted Q control signal 2596.

[0800] In a preferred embodiment, control signals 2590,2592,2594 and2596 output pulses having a width equal to one-half of a period of I/Qmodulated RF input signal 2582. The invention, however, is not limitedto these pulse widths, and control signals 2590, 2592, 2594, and 2596may comprise pulse widths of any fraction of, or multiple and fractionof, a period of I/Q modulated RF input signal 2582.

[0801] First, second, third, and fourth pulse generators 2610, 2612,2614, and 2616 may be implemented in circuit logic, hardware, software,or any combination thereof, as would be known by persons skilled in therelevant art(s).

[0802] As shown in FIG. 27, control signals 2590, 2592,2594, and 2596comprise pulses that are non-overlapping. Furthermore, in this example,pulses appear on these signals in the following order: I control signal2590, Q control signal 2594, inverted I control signal 2592, andinverted Q control signal 2596. Potential circuit re-radiation from I/Qmodulation receiver 2500 may comprise frequency components from acombination of these control signals.

[0803] For example, FIG. 28 shows an overlay of pulses from I controlsignal 2590, Q control signal 2594, inverted I control signal 2592, andinverted Q control signal 2596. When pulses from these control signalsleak to through first, second, third, and fourth UFD modules 2502,2506,2510, and 2514 of to antenna 2582 (shown in FIG. 25), they may beradiated from I/Q modulation receiver 2500, with a combined waveformthat appears to have a primary frequency equal to four times thefrequency of any single one of control signals 2590, 2592, 2594, and2596. FIG. 27 shows an example combined control signal 2702.

[0804]FIG. 28 also shows an example I/Q modulation RF input signal 2582overlaid upon control signals 2590, 2592, 2594, and 2596. As shown inFIG. 28, pulses on I control signal 2590 overlay and act to down-converta positive I-phase portion of I/Q modulation RF input signal 2582.Pulses on inverted I control signal 2592 overlay and act to down-converta negative I-phase portion of I/Q modulation RF input signal 2582.Pulses on Q control signal 2594 overlay and act to down-convert a risingQ-phase portion of I/Q modulation RF input signal 2582. Pulses oninverted Q control signal 2596 overlay and act to down-convert a fallingQ-phase portion of I/Q modulation RF input signal 2582.

[0805] As FIG. 28 further shows in this example, the frequency ratiobetween the combination of control signals 2590, 2592, 2594, and 2596and I/Q modulation RF input signal 2582 is 4:3. Because the frequency ofthe potentially re-radiated signal, combined control signal 2702, issubstantially different from that of the signal being down-converted,I/Q modulation RF input signal 2582, it does not interfere with signaldown-conversion as it is out of the frequency band of interest, andhence may be filtered out. In this manner, I/Q modulation receiver 2500reduces problems due to circuit re-radiation. As will be understood bypersons skilled in the relevant art(s) from the teachings herein,frequency ratios other than 4:3 may be implemented to achieve similarreduction of problems of circuit re-radiation.

[0806] It should be understood that the above control signal generatorcircuit example is provided for illustrative purposes only. Theinvention is not limited to these embodiments. Alternative embodiments(including equivalents, extensions, variations, deviations, etc., of theembodiments described herein) for I/Q modulation control signalgenerator 2600 will be apparent to persons skilled in the relevantart(s) from the teachings herein, and are within the scope of thepresent invention.

[0807] 7.4.1.2 Detailed Example I/Q Modulation Receiver Embodiment withExemplary Waveforms

[0808]FIG. 29 illustrates a more detailed example circuit implementationof I/Q modulation receiver 2500, according to an embodiment of thepresent invention.

[0809] FIGS. 30-40 show waveforms related to an example implementationof I/Q modulation receiver 2500 of FIG. 29.

[0810]FIGS. 30 and 31 show first and second input data signals 2902 and2904 to be I/Q modulated with a RF carrier signal frequency as theI-phase and Q-phase information signals, respectively.

[0811]FIGS. 33 and 34 show the signals of FIGS. 30 and 31 aftermodulation with a RF carrier signal frequency, respectively, asI-modulated signal 2906 and Q-modulated signal 2908.

[0812]FIG. 32 shows an I/Q modulation RF input signal 2582 formed fromI-modulated signal 2906 and Q-modulated signal 2908 of FIGS. 33 and 34,respectively.

[0813]FIG. 39 shows an overlaid view of filtered I output signal 3902and filtered inverted I output signal 3904.

[0814]FIG. 40 shows an overlaid view of filtered Q output signal 4002and filtered inverted Q output signal 4004.

[0815]FIGS. 35 and 36 show I baseband output signal 2584 and Q basebandoutput signal 2586, respectfully. A data transition 3202 is indicated inboth I baseband output signal 2584 and Q baseband output signal 2586.The corresponding data transition 3202 is indicated in I-modulatedsignal 2906 of FIG. 33, Q-modulated signal 2908 of FIG. 34, and I/Qmodulation RF input signal 2582 of FIG. 32.

[0816]FIGS. 37 and 38 show I baseband output signal 2584 and Q basebandoutput signal 2586 over a wider time interval.

[0817] 7.4.1.3 Example Single Channel Receiver Embodiment

[0818]FIG. 41 illustrates an exemplary single channel receiver 4100,corresponding to either the I or Q channel of I/Q modulation receiver2500, according to an embodiment of the present invention. Singlechannel receiver 4100 can down-convert an input RF signal 4106 modulatedaccording to AM, PM, FM, and other modulation schemes. Refer to section7.4.1 above for further description on the operation of single channelreceiver 4100.

[0819] 7.4.1.4 Alternative Example I/Q Modulation Receiver Embodiment

[0820]FIG. 42 illustrates an exemplary I/Q modulation receiver 4200,according to an embodiment of the present invention. I/Q modulationreceiver 4200 receives, down-converts, and demodulates an I/Q modulatedRF input signal 2582 to an I baseband output signal 2584, and a Qbaseband output signal 2586. I/Q modulation receiver 4200 has additionaladvantages of reducing or eliminating unwanted DC offsets and circuitre-radiation, in a similar fashion to that of I/Q modulation receiver2500 described above.

[0821] 7.4.1.5 Example Transmitter Embodiment

[0822]FIG. 43 illustrates an exemplary I/Q modulation transmitter 4300(only I channel is shown), according to an embodiment of the presentinvention. I/Q modulation transmitter has a configuration substantiallysimilar to I/Q modulation receiver 2500. Hence, an I/Q modulationtransmitter 4300 and an I/Q modulation receiver 2500 may be implementedwith at least some common circuit components.

[0823] I/Q modulation transmitter 4300 comprises an optional firstfilter 4302, a second optional filter 4306, and a third optional filter4310. When present, second and third optional filters 4306 and 4310 maycomprise first and second resistors 4334 and 4336, respectively. Inalternative embodiments, second and third optional filters 4306 and 4310may comprise inductors, capacitors, and/or other filtering elements,alone or in combination.

[0824] 8. Conclusion

[0825] While various embodiments of the present invention have beendescribed above, it should be understood that they have been presentedby way of example only, and not limitation. It will be apparent topersons skilled in the relevant art that various changes in form anddetail can be made therein without departing from the spirit and scopeof the invention. Thus, the breadth and scope of the present inventionshould not be limited by any of the above-described exemplaryembodiments, but should be defined only in accordance with the followingclaims and their equivalents.

What is claimed is:
 1. An apparatus for down-converting anelectromagnetic signal and reducing DC offset voltages and re-radiation,comprising: a first UFD module that receives an input signal, whereinsaid first UFD module down-converts said input signal according to afirst control signal and outputs a first down-converted signal; a secondUFD module that receives said input signal, wherein said second UFDmodule down-converts said input signal according to a second controlsignal and outputs a second down-converted signal; and a firstsubtractor module that subtracts said second down-converted signal fromsaid first down-converted signal and outputs a first channeldown-converted signal.